Backhaul radio with a substrate tab-fed antenna assembly

ABSTRACT

Directive gain antenna elements implemented with an aperture-fed patch array antenna assembly are described. A feed network for the aperture-fed patch array may include offset apertures and may also include meandering feed lines. Scalable aperture shapes and orientations that can be used with antennas operating at any frequency and with dual orthogonal polarizations are also disclosed. Directive gain antenna elements implemented with arrays of orthogonal reflected dipoles are also described with optimal feed networks and parasitic elements to achieve desired directive gain characteristics. Such arrayed dipole antennas feature dual orthogonal polarizations with assembly tabs that lower cost and improve reliability. Backhaul radios that incorporate said antennas are also disclosed.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a continuation application of U.S. patentapplication Ser. No. 14/197,158, filed on Mar. 4, 2014, which claimspriority to U.S. patent application Ser. No. 13/645,472, filed on Oct.4, 2012, which claims priority to U.S. Pat. No. 8,311,023 and U.S. Pat.No. 8,238,318, the disclosures of which are hereby incorporated hereinby reference in their entirety.

The present application is also related to U.S. patent application Ser.No. 13/898,429, filed May 20, 2013 and U.S. Pat. No. 8,467,363, thedisclosures of which are hereby incorporated herein by reference intheir entirety.

The present application is also related to U.S. patent application Ser.No. 13/271,051, filed Oct. 11, 2011 and U.S. Pat. No. 8,300,590, thedisclosures of which are hereby incorporated herein by reference intheir entirety.

The present application is also related to U.S. patent application Ser.No. 14/108,200, filed Dec. 16, 2013 and U.S. Pat. Nos. 8,638,839 and8,422,540, the disclosures of which are hereby incorporated herein byreference in their entirety.

BACKGROUND

1. Field

The present disclosure relates generally to data networking and inparticular to a backhaul radio for connecting remote edge accessnetworks to core networks.

2. Related Art

Data networking traffic has grown at approximately 100% per year forover 20 years and continues to grow at this pace. Only transport overoptical fiber has shown the ability to keep pace with thisever-increasing data networking demand for core data networks. Whiledeployment of optical fiber to an edge of the core data network would beadvantageous from a network performance perspective, it is oftenimpractical to connect all high bandwidth data networking points withoptical fiber at all times. Instead, connections to remote edge accessnetworks from core networks are often achieved with wireless radio,wireless infrared, and/or copper wireline technologies.

Radio, especially in the form of cellular or wireless local area network(WLAN) technologies, is particularly advantageous for supportingmobility of data networking devices. However, cellular base stations orWLAN access points inevitably become very high data bandwidth demandpoints that require continuous connectivity to an optical fiber corenetwork.

When data aggregation points, such as cellular base station sites, WLANaccess points, or other local area network (LAN) gateways, cannot bedirectly connected to a core optical fiber network, then an alternativeconnection, using, for example, wireless radio or copper wirelinetechnologies, must be used. Such connections are commonly referred to as“backhaul.”

Many cellular base stations deployed to date have used copper wirelinebackhaul technologies such as T1, E1, DSL, etc. when optical fiber isnot available at a given site. However, the recent generations of HSPA+and LTE cellular base stations have backhaul requirements of 100 Mb/s ormore, especially when multiple sectors and/or multiple mobile networkoperators per cell site are considered. WLAN access points commonly havesimilar data backhaul requirements. These backhaul requirements cannotbe practically satisfied at ranges of 300 m or more by existing copperwireline technologies. Even if LAN technologies such as Ethernet overmultiple dedicated twisted pair wiring or hybrid fiber/coax technologiessuch as cable modems are considered, it is impractical to backhaul atsuch data rates at these ranges (or at least without adding intermediaterepeater equipment). Moreover, to the extent that such special wiring(i.e., CAT 5/6 or coax) is not presently available at a remote edgeaccess network location; a new high capacity optical fiber isadvantageously installed instead of a new copper connection.

Rather than incur the large initial expense and time delay associatedwith bringing optical fiber to every new location, it has been common tobackhaul cell sites, WLAN hotspots, or LAN gateways from offices,campuses, etc. using microwave radios. An exemplary backhaul connectionusing the microwave radios 132 is shown in FIG. 1. Traditionally, suchmicrowave radios 132 for backhaul have been mounted on high towers 112(or high rooftops of multi-story buildings) as shown in FIG. 1, suchthat each microwave radio 132 has an unobstructed line of sight (LOS)136 to the other. These microwave radios 132 can have data rates of 100Mb/s or higher at unobstructed LOS ranges of 300 m or longer withlatencies of 5 ms or less (to minimize overall network latency).

Traditional microwave backhaul radios 132 operate in a Point to Point(PTP) configuration using a single “high gain” (typically >30 dBi oreven >40 dBi) antenna at each end of the link 136, such as, for example,antennas constructed using a parabolic dish. Such high gain antennasmitigate the effects of unwanted multipath self-interference or unwantedco-channel interference from other radio systems such that high datarates, long range and low latency can be achieved. These high gainantennas however have narrow radiation patterns.

Furthermore, high gain antennas in traditional microwave backhaul radios132 require very precise, and usually manual, physical alignment oftheir narrow radiation patterns in order to achieve such highperformance results. Such alignment is almost impossible to maintainover extended periods of time unless the two radios have a clearunobstructed line of sight (LOS) between them over the entire range ofseparation. Furthermore, such precise alignment makes it impractical forany one such microwave backhaul radio to communicate effectively withmultiple other radios simultaneously (i.e., a “point to multipoint”(PMP) configuration).

In wireless edge access applications, such as cellular or WLAN, advancedprotocols, modulation, encoding and spatial processing across multipleradio antennas have enabled increased data rates and ranges for numeroussimultaneous users compared to analogous systems deployed 5 or 10 yearsago for obstructed LOS propagation environments where multipath andco-channel interference were present. In such systems, “low gain”(usually <6 dBi) antennas are generally used at one or both ends of theradio link both to advantageously exploit multipath signals in theobstructed LOS environment and allow operation in different physicalorientations as would be encountered with mobile devices. Althoughimpressive performance results have been achieved for edge access, suchresults are generally inadequate for emerging backhaul requirements ofdata rates of 100 Mb/s or higher, ranges of 300 m or longer inobstructed LOS conditions, and latencies of 5 ms or less.

In particular, “street level” deployment of cellular base stations, WLANaccess points or LAN gateways (e.g., deployment at street lamps, trafficlights, sides or rooftops of single or low-multiple story buildings)suffers from problems because there are significant obstructions for LOSin urban environments (e.g., tall buildings, or any environments wheretall trees or uneven topography are present).

FIG. 1 illustrates edge access using conventional unobstructed LOS PTPmicrowave radios 132. The scenario depicted in FIG. 1 is common for many2^(nd) Generation (2G) and 3^(rd) Generation (3G) cellular networkdeployments using “macrocells”. In FIG. 1, a Cellular Base TransceiverStation (BTS) 104 is shown housed within a small building 108 adjacentto a large tower 112. The cellular antennas 116 that communicate withvarious cellular subscriber devices 120 are mounted on the towers 112.The PTP microwave radios 132 are mounted on the towers 112 and areconnected to the BTSs 104 via an nT1 interface. As shown in FIG. 1 byline 136, the radios 132 require unobstructed LOS.

The BTS on the right 104 a has either an nT1 copper interface or anoptical fiber interface 124 to connect the BTS 104 a to the Base StationController (BSC) 128. The BSC 128 either is part of or communicates withthe core network of the cellular network operator. The BTS on the left104 b is identical to the BTS on the right 104 a in FIG. 1 except thatthe BTS on the left 104 b has no local wireline nT1 (or optical fiberequivalent) so the nT1 interface is instead connected to a conventionalPTP microwave radio 132 with unobstructed LOS to the tower on the right112 a. The nT1 interfaces for both BTSs 104 a, 104 b can then bebackhauled to the BSC 128 as shown in FIG. 1.

In the conventional PTP radios 132, as described in greater detail inU.S. patent application Ser. No. 13/645,472 and incorporated herein, theantenna is typically of very high gain such as can be achieved by aparabolic dish so that gains of typically >30 dBi (or even sometimes >40dBi), can be realized. Such an antenna usually has a narrow radiationpattern in both the elevation and azimuth directions. The use of such ahighly directive antenna in a conventional PTP radio link withunobstructed LOS propagation conditions ensures that a modem within suchradios has insignificant impairments at the receiver due to multipathself-interference and further substantially reduces the likelihood ofunwanted co-channel interference due to other nearby radio links.However, the conventional PTP radio on a whole is completely unsuitablefor obstructed LOS or PMP operation.

In U.S. patent application Ser. No. 13/645,472 and the relatedapplications and patents summarized above, a novel Intelligent BackhaulRadio (or “IBR”) suitable for obstructed LOS and PMP or PTP operation isdescribed in great detail in various embodiments of those inventions.Additionally, in U.S. patent application Ser. No. 13/898,429, certainexemplary antenna assemblies were described. Applicants have identifiedherein additional improvements to antenna assembly designs for bothpatch-based and dipole-based radiating element structures.

Aperture-fed antennas have been previously known in the art. Forexample, in D. M. Pozar, “A microstrip antenna aperture-coupled to amicrostripline,” Electron. Lett., vol. 21, no. 2, pp. 49-50, 1985, andin D. M. Pozar and S. D. Targonski, “Improved coupling foraperture-coupled microstrip antennas,” Electron. Lett., vol. 27, no. 13,pp. 1129-1131, 1991, an aperture-fed patch antenna was disclosed.Additionally, in S. C. Gao et al., “Dual-polarized slot-coupled planarantenna with wide bandwidth,” IEEE Trans. Antennas and Propagation, vol.51, no. 3, pp. 441-448, 2003, a dual-polarization aperture-fed antennawas disclosed. However, the conventional art is completely unsuitablefor application in an IBR. For example, the conventional aperture fedantennas have insufficient antenna gain for IBR directive gain antennaelements, have unacceptable coupling efficiencies, have unacceptablebackwards facing radiation and are impractical to manufacturecost-effectively and reliably.

SUMMARY

The following summary of the invention is included in order to provide abasic understanding of some aspects and features of the invention. Thissummary is not an extensive overview of the invention and as such it isnot intended to particularly identify key or critical elements of theinvention or to delineate the scope of the invention. Its sole purposeis to present some concepts of the invention in a simplified form as aprelude to the more detailed description that is presented below.

Some embodiments of the claimed inventions are directed to an improvedantenna assembly including an array of resonant radiating patch antennaelements and transmission line feed networks that areelectromagnetically coupled using apertures. Other embodiments of theclaimed inventions are directed to an improved antenna assemblyincluding an array of dipole antenna elements and transmission line feednetworks that are conductively connected at junctions formed withsubstrate tabs and cutouts. Backhaul radios that include the improvedantenna assemblies are also disclosed.

According to an aspect of the invention, an antenna assembly is providedthat includes a first substrate comprising a plurality of conductivepatch elements; a second substrate comprising a first layer with atleast a conductive ground plane and a plurality of pairs of apertures,wherein the number of pairs of apertures is equal to the number ofconductive patch elements, and a second layer with at least a firsttransmission line feed network coupled to a first feed point and asecond transmission line feed network coupled to a second feed point;and a spacer interposed between the first substrate and the secondsubstrate, the spacer comprising a dielectric material and at least onespacer opening in the dielectric material, wherein the dielectricmaterial is absent within the at least one spacer opening; wherein thefirst transmission line feed network overlaps a first aperture of eachpair of the plurality of pairs of apertures and the second transmissionline feed network overlaps a second aperture of each pair of theplurality of pairs of apertures; wherein the first aperture of each pairof the plurality of pairs of apertures electromagnetically couples thefirst transmission line feed network and the second aperture of eachpair of the plurality of pairs of apertures electromagnetically couplesthe second transmission line feed network to a respective one of theplurality of conductive patch elements; and wherein the first apertureof each pair of the plurality of pairs of apertures is orthogonal to thesecond aperture of each pair of the plurality of pairs of apertures.

The first substrate may be a printed circuit board. The second substratemay be a printed circuit board. The second substrate may be a printedcircuit board having more than two layers.

The first transmission line feed network and the second transmissionline feed network each may include striplines. The first transmissionline feed network and the second transmission line feed network each mayinclude microstrip lines. The first feed point and the second feed pointmay each be coupled to respective components on an outside layer of thesecond substrate. The respective components may be at least one of an RFbandpass filter or a low noise amplifier within a receiver.

The at least one spacer opening may extend beyond a projected area ofone or more of the plurality of conductive patch elements by at least adistance equal to a thickness of the spacer.

The first aperture of each pair of the plurality of pairs of aperturesmay excite a respective resonant radiating cavity formed between eachrespective one of the plurality of conductive patch elements and theconductive ground plane in an electromagnetic mode corresponding to avertical polarization far-field pattern, and wherein the second apertureof each pair of the plurality of pairs of apertures may excite saidrespective resonant radiating cavity in an electromagnetic modecorresponding to a horizontal polarization far-field pattern.

The antenna assembly may further include a plurality of plasticfasteners to hold the first substrate, the second substrate and thespacer together.

The first aperture of each respective pair of the plurality of pairs ofapertures may be oriented relative to the second aperture of eachrespective pair of the plurality of pairs of apertures in a T-shape.Each of the first aperture and the second aperture of each respectivepair of the plurality of pairs of apertures may include a rectangularaperture body with an aperture body width and a pair of aperture endswith an aperture end width. Each aperture end may include a rectangularend and a semi-circular end with a radius equal to one half of theaperture end width. The aperture end width may be at least five timesgreater than the aperture body width. Each aperture end may be taperedor rounded. The rectangular end may have a width equal to the apertureend width and a thickness equal to one sixth of the aperture end width.The aperture end width may be equal to one third of an aperture length.

The first transmission line feed network may be terminated by a firstvia to the conductive ground plane after a feedline portion of the firsttransmission line feed network crosses over the rectangular aperturebody of the first aperture of each pair of the plurality of pairs ofapertures, and the second transmission line feed network may beterminated by a second via to the conductive ground plane after afeedline portion of the second transmission line feed network crossesover the rectangular aperture body of the second aperture of each pairof the plurality of pairs of apertures.

The plurality of conductive patch elements may be arranged in an arraywith a plurality of rows wherein each row comprises at least oneconductive patch element. The plurality of conductive patch elements maybe arranged in an array with a plurality of rows and a plurality ofcolumns wherein each row comprises a number of conductive patch elementsequal to the number of columns. The number of columns may be equal totwo.

A first feedline portion of the first transmission line feed network maycross over a rectangular aperture body of the first aperture of eachpair of the plurality of pairs of apertures in a first direction foreach first aperture that excites each respective resonant radiatingcavity formed between each respective one of the plurality of conductivepatch elements and the conductive ground plane for conductive patchelements may be arranged in a first column and a second feedline portionof the first transmission line feed network may cross over therectangular aperture body of the first aperture of each pair of theplurality of pairs of apertures in a second direction for each firstaperture that excites each respective resonant radiating cavity formedbetween each respective one of the plurality of conductive patchelements and the conductive ground plane for conductive patch elementsarranged in a second column, and the second direction may be opposite tothe first direction.

The second feedline portion may be electrically longer than the firstfeedline portion by a distance equivalent to 180 degrees in phase at atarget operating frequency for the antenna assembly.

A third feedline portion of the second transmission line feed networkmay cross over a rectangular aperture body of the second aperture ofeach pair of the plurality of pairs of apertures in a third directionfor each second aperture that excites each respective resonant radiatingcavity formed between each respective one of the plurality of conductivepatch elements and the conductive ground plane for conductive patchelements arranged in the first column and a fourth feedline portion ofthe second transmission line feed network may cross over the rectangularaperture body of the second aperture of each pair of the plurality ofpairs of apertures in a fourth direction for each second aperture thatexcites each respective resonant radiating cavity formed between eachrespective one of the plurality of conductive patch elements and theconductive ground plane for conductive patch elements arranged in thesecond column, and the third direction may be the same as the fourthdirection.

The third feedline portion may be equivalent in electrical length to thefourth feedline portion. Each of the first transmission line feednetwork and the second transmission line feed network may include atleast one meandering line portion. Each meandering line portion mayinclude one or more bends, and wherein an electrical length of eachmeandering line portion may match a group delay from the respectivefirst or second feed point to at least one of the respective first orsecond apertures with that of another group delay from the respectivefirst or second feed point to at least one other of the respective firstor second apertures.

Each of the first transmission line feed network and the secondtransmission line feed network may include at least one tunable element.An input signal applied to at least one tunable element may adjust atleast one characteristic of the antenna assembly, said characteristicbeing at least one selected from the group consisting of a far-fieldradiation pattern, a coupling between the first feed point and thesecond feed point, and a coupling to one or more nearby antennas.

According to another aspect of the invention, an antenna assembly isprovided that includes a plurality of first substrates each comprising aunitary dipole antenna element, wherein each unitary dipole antennaelement comprises a first pair of dipole branches, a first coplanar feedline pair and a first conductor connection substrate tab; a secondsubstrate comprising a plurality of coplanar dipole antenna elements,wherein each coplanar dipole antenna element comprises a second pair ofdipole branches, a second coplanar feed line pair and a second conductorconnection substrate tab; and a third substrate comprising a pluralityof conductor connection cutouts, a first layer and a second layer,wherein the first layer comprises a conductive plane with a plurality ofconductor connection clearances and wherein the second layer comprises afirst transmission line feed network and a second transmission line feednetwork; wherein the second substrate is orthogonal to each of theplurality of first substrates and wherein the third substrate isorthogonal to the second substrate and each of the plurality of firstsubstrates; wherein the first transmission line feed networkconductively connects to each respective unitary dipole antenna elementvia its respective first coplanar feed line pair at a respective one ofa plurality of first conductive junctions, each said first conductivejunction comprising the respective first conductor connection substratetab, a first corresponding one of the plurality of conductor connectioncutouts, and a first corresponding one of the plurality of conductorconnection clearances; and wherein the second transmission line feednetwork conductively connects to each respective coplanar dipole antennaelement via its respective second coplanar feed line pair at arespective one of a plurality of second conductive junctions, each saidsecond conductive junction comprising the respective second conductorconnection substrate tab, a second corresponding one of the plurality ofconductor connection cutouts, and a second corresponding one of theplurality of conductor connection clearances.

The first pair of dipole branches of each unitary dipole antenna elementmay be located on a same surface as the first coplanar feed line pair.Each unitary dipole antenna element may further include a first pair ofparasitic elements. The first pair of parasitic elements of each unitarydipole antenna element may be located on the same surface as the firstpair of dipole branches.

The first pair of parasitic elements may broaden a radiation pattern ofeach unitary dipole antenna element in a plane of the same surface asthe first pair of dipole branches. The first pair of parasitic elementsmay include half-wavelength resonant dipole elements at a targetoperating frequency of the antenna assembly. The first pair of parasiticelements may be asymmetrically offset relative to an axis of therespective first pair of dipole branches towards an end of therespective first substrate having the respective first conductorconnection substrate tab.

Each of the plurality of first substrates may further include a firstassembly slot and the second substrate may further include a pluralityof second assembly slots. A respective one of the plurality of secondassembly slots may align with a respective first assembly slot withineach respective first substrate.

Each of the plurality of first substrates further include one or morefirst mechanical tabs. The third substrate may further includeadditional cutouts, each additional cutout corresponding to a respectivefirst mechanical tab amongst the plurality of first substrates.

Each of the plurality of first substrates may further include one ormore first metalized pads corresponding to respective ones of each firstmechanical tab. The second layer of the third substrate may furtherinclude a plurality of third metalized pads corresponding to respectiveones of each first mechanical tab. Each first metalized pad may adjoin arespective third metalized pad.

The second substrate may further include one or more second mechanicaltabs. The third substrate may further include additional cutouts, eachadditional cutout corresponding to a respective second mechanical tab.

The second substrate may further include one or more second metalizedpads corresponding to respective ones of each second mechanical tab. Thesecond layer of the third substrate may further include a plurality ofthird metalized pads corresponding to respective ones of each secondmechanical tab. Each second metalized pad may adjoin a respective thirdmetalized pad.

Each of the plurality of conductor connection clearances may beasymmetrically offset relative to a respective one of the plurality ofconductor connection cutouts. The asymmetric offset may center each ofthe plurality of conductor connection clearances relative to a projectedintersection with the third substrate for a respective one of firstcoplanar feed line pairs or second coplanar feed line pairs.

The second substrate may be oriented such that each of the plurality ofcoplanar dipole antenna elements radiates in a vertical polarizationfar-field pattern and the plurality of first substrates may be orientedsuch that each unitary dipole antenna element radiates in a horizontalpolarization far-field pattern.

The first transmission line feed network may include a first feed point,a first microstrip distribution portion, and a plurality of firstmicrostrip feed structure portions and the second transmission line feednetwork may include a second feed point, a second microstripdistribution portion, and a plurality of second microstrip feedstructure portions.

Each first microstrip feed structure portion may include a first balunstructure that couples a first pair of balanced microstrip lines at arespective one of the plurality of first conductive junctions to a firstunbalanced microstrip line within the first microstrip distributionportion and each second microstrip feed structure portion may include asecond balun structure that couples a second pair of balanced microstriplines at a respective one of the plurality of second conductivejunctions to a second unbalanced microstrip line within the secondmicrostrip distribution portion.

Each of the first and second balun structures may include a firstmicrostrip line, a second microstrip line, and a T-junction, and thesecond microstrip line may be electrically longer than the firstmicrostrip line by one half wavelength at a target operating frequencyof the antenna assembly and the second microstrip line may include atleast one additional bend than the first microstrip line.

Each of the first and second microstrip lines may function as animpedance transformer of an electrical length that is an integermultiple of one quarter wavelength at a target operating frequency ofthe antenna assembly.

Each of the first microstrip feed structure portion and the secondmicrostrip feed structure portion may further include an impedancetransformer from the T-junction within its respective first or secondbalun structure to its respective first or second unbalanced microstripline within the respective first or second microstrip distributionportion. The impedance transformer may include an unbalanced microstripline of an electrical length that is an integer multiple of one quarterwavelength at a target operating frequency of the antenna assembly.

The first feed point and the second feed point may each be coupled torespective components on the second layer of the third substrate. Therespective components may be at least one of an RF filter or a poweramplifier within a transmitter.

The first microstrip distribution portion may equally divide a firstpower and matches a first group delay from the first feed point to eachof the plurality of first microstrip feed structure portions and thesecond microstrip distribution portion may equally divide a second powerand matches a second group delay from the second feed point to each ofthe plurality of second microstrip feed structure portions. Each of thefirst microstrip distribution portion and the second microstripdistribution portion may include at least one tunable element. An inputsignal applied to at least one tunable element may adjust at least onecharacteristic of the antenna assembly, said characteristic being one ormore of a far-field radiation pattern, a coupling between the first feedpoint and the second feed point, or a coupling to one or more nearbyantennas.

A numerical count of unitary dipole antenna elements may exceed that ofa numerical count of coplanar dipole antenna elements.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are incorporated into and constitute apart of this specification, illustrate one or more examples ofembodiments and, together with the description of example embodiments,serve to explain the principles and implementations of the embodiments.

FIG. 1 is an illustration of conventional point to point (PTP) radiosdeployed for cellular base station backhaul with unobstructed line ofsight (LOS).

FIG. 2 is an illustration of intelligent backhaul radios (IBRs) deployedfor cellular base station backhaul with obstructed LOS according to oneembodiment of the invention.

FIG. 3 is a block diagram of an IBR according to one embodiment of theinvention.

FIG. 4 is a block diagram of an IBR antenna array according to oneembodiment of the invention.

FIG. 5A is an assembly view of an antenna assembly according to oneembodiment of the invention.

FIG. 5B is a side view of the antenna assembly according to oneembodiment of the invention.

FIG. 5C is an assembly view of an alternate embodiment of the invention.

FIG. 6 is a view of the plurality of conductive patch elements on thefirst substrate of the antenna assembly according to one embodiment ofthe invention.

FIG. 7 is a view of the spacer laid over the plurality of conductivepatch elements on the first substrate of the antenna assembly accordingto one embodiment of the invention.

FIG. 8A is a detailed view of both the first layer and the second layerof the second substrate of the antenna assembly according to oneembodiment of the invention.

FIG. 8B is a detailed view of the transmission line feed networkportions near the apertures of the second substrate of the antennaassembly according to one embodiment of the invention.

FIG. 8C is a detailed view of the first layer of the second substrate ofthe antenna assembly according to one embodiment of the invention.

FIG. 8D is a detailed view of the second layer of the second substrateof the antenna assembly according to one embodiment of the invention.

FIG. 8E is a detailed view of one of the plurality of apertures withinthe first layer of the second substrate of the antenna assemblyaccording to one embodiment of the invention.

FIG. 8F is a view of the first and second substrates showing how theplurality of pairs of apertures on the first layer of the secondsubstrate align with the plurality of conductive patch elements on thefirst substrate according to one embodiment of the invention.

FIG. 9 is a view showing the surface current of the second substrateusing the aperture feed arrangement according to one embodiment of theinvention.

FIG. 10A is a detailed view of a unitary dipole antenna element for adipole array antenna assembly according to one embodiment of theinvention.

FIG. 10B is a detailed view of a plurality of coplanar dipole antennaelements for a dipole array antenna assembly according to one embodimentof the invention.

FIG. 11A is a detailed view of a microstrip feed structure portion for adipole array antenna assembly according to one embodiment of theinvention.

FIG. 11B is a detailed view of an orthogonal interconnection ofsubstrates for a dipole array antenna assembly according to oneembodiment of the invention.

FIG. 12 is a schematic diagram of cascade impedances for a dipoleantenna array assembly according to one embodiment of the invention.

FIG. 13A is an assembly view of a dipole array antenna assemblyaccording to one embodiment of the invention.

FIG. 13B is an alternative assembly view of a dipole array antennaassembly according to one embodiment of the invention.

FIG. 14 is a detailed view of first and second layers of the thirdsubstrate of a dipole array antenna assembly according to one embodimentof the invention.

DETAILED DESCRIPTION

FIG. 2 illustrates deployment of intelligent backhaul radios (IBRs) inaccordance with an embodiment of the invention. As shown in FIG. 2, theIBRs 200 are deployable at street level with obstructions such as trees204, hills 208, buildings 212, etc. between them. The IBRs 200 are alsodeployable in configurations that include point to multipoint (PMP), asshown in FIG. 2, as well as point to point (PTP). In other words, eachIBR 200 may communicate with more than one other IBR 200.

For 3G and especially for 4^(th) Generation (4G), cellular networkinfrastructure is more commonly deployed using “microcells” or“picocells.” In this cellular network infrastructure, compact basestations (eNodeBs) 216 are situated outdoors at street level. When sucheNodeBs 216 are unable to connect locally to optical fiber or a copperwireline of sufficient data bandwidth, then a wireless connection to afiber “point of presence” (POP) requires obstructed LOS capabilities, asdescribed herein.

For example, as shown in FIG. 2, the IBRs 200 include an Aggregation EndIBR (AE-IBR) and Remote End IBRs (RE-IBRs). The eNodeB 216 associatedwith the AE-IBR is typically connected locally to the core network via afiber POP 220. The RE-IBRs and their associated eNodeBs 216 aretypically not connected to the core network via a wireline connection;instead, the RE-IBRs are wirelessly connected to the core network viathe AE-IBR. As shown in FIG. 2, the wireless connections between theIBRs include obstructions (i.e., there may be an obstructed LOSconnection between the RE-IBRs and the AE-IBR).

FIG. 3 illustrates an exemplary embodiment of the IBRs 200 shown in FIG.2. In FIG. 3, the IBRs 200 include interfaces 304, interface bridge 308,MAC 312, modem 324, channel MUX 328, RF 332, which includes Tx1 . . .TxM 336 and Rx1 . . . RxN 340, antenna array 348 (includes multipleantennas 352), a Radio Link Controller (RLC) 356 and a Radio ResourceController (RRC) 360. The IBR may optionally include an IntelligentBackhaul Management System (IBMS) agent as shown in FIG. 7 of U.S.patent application Ser. No. 13/645,472. It will be appreciated that thecomponents and elements of the IBRs may vary from that illustrated inFIG. 3. U.S. patent application Ser. No. 13/645,472 and the relatedapplications and patents summarized above describe in detail the variouselements of the IBR including their structural and operational featuresin numerous different embodiments both as depicted in FIG. 3 and asdepicted with various additional elements not shown in FIG. 3. A briefsummary of certain elements of the IBR is also provided herein.

The external interfaces of the IBR (i.e., the IBR Interface Bridge 308on the wireline side and the IBR Antenna Array 348 (including antennas352) on the wireless side) are a starting point for describing somefundamental differences between the numerous different embodiments ofthe IBR 200 and either conventional PTP radios or other commonly knownradio systems, such as those built to existing standards including802.11n (WiFi), 802.11ac (WiFi), 802.16e (WiMax) or 4G LTE.

In some embodiments, the IBR Interface Bridge 308 physically interfacesto standards-based wired data networking interfaces 304 as Ethernet 1through Ethernet P. “P” represents a number of separate Ethernetinterfaces over twisted-pair, coax or optical fiber. The IBR InterfaceBridge 308 can multiplex and buffer the P Ethernet interfaces 304 withthe IBR MAC 312. In exemplary embodiments, the IBR Interface Bridge 308preserves “Quality of Service” (QoS) or “Class of Service” (CoS)prioritization as indicated, for example, in IEEE 802.1q 3-bit PriorityCode Point (PCP) fields within the Ethernet frame headers, such thateither the IBR MAC 312 schedules such frames for transmission accordingto policies configured within or communicated to the IBR 200, or the IBRinterface bridge 308 schedules the transfer of such frames to the IBRMAC 312 such that the same net effect occurs. In other embodiments, theIBR interface bridge 308 also forwards and prioritizes the delivery offrames to or from another IBR over an instant radio link based onMultiprotocol Label Switching (MPLS) or Multiprotocol Label SwitchingTransport Profile (MPLS-TP). U.S. patent application Ser. No. 13/645,472provides additional description of exemplary embodiments of theinterfaces 304 and the interface bridge 308 of the IBR 200. U.S. patentapplication Ser. No. 13/271,051 provides additional description ofexemplary embodiments of an IBMS that includes an IBMS Agent incommunication with or IBMS components and the IBR Interface Bridge 308as well as MAC 312 and/or RRC 360. U.S. patent application Ser. No.13/271,051 also describes an IBR with an integrated Carrier Ethernetswitch.

FIG. 4 illustrates an exemplary embodiment of an IBR Antenna Array 348.FIG. 4 illustrates an antenna array having Q directive gain antennas 352(i.e., where the number of antennas is greater than 1). In FIG. 4, theIBR Antenna Array 348 includes an IBR RF Switch Fabric 412, RFinterconnections 404, a set of Front-ends 408 and the directive gainantennas 352. The RF interconnections 404 can be, for example, circuitboard traces and/or coaxial cables. The RF interconnections 404 connectthe IBR RF Switch Fabric 412 and the set of Front-ends 408. EachFront-end 408 is associated with an individual directive gain antenna352, numbered consecutively from 1 to Q.

U.S. patent application Ser. No. 13/645,472, U.S. patent applicationSer. No. 13/898,429, and U.S. patent application Ser. No. 14/108,200provide additional description of the Front-end 408 and variousembodiments thereof as applicable to different IBR duplexing schemessuch as Time Division Duplexing (TDD), Frequency Division Duplexing(FDD) and Zero Division Duplexing (ZDD). For example, with TDDembodiments where certain directive gain antenna elements 352 are usedfor both transmit and receive at different times, then Front-end 408 mayinclude a transmit/receive switch, one or more RF low pass and/orbandpass filters, and either a low-noise amplifier (LNA) in the receivepath or a power amplifier (PA) in the transmit path. Similarly, with FDDembodiments where certain directive gain antenna elements 352 are usedfor both transmit and receive at the same time, then Front-end 408 mayinclude a duplex filter, one or more additional RF low pass and/orbandpass filters, and either a low-noise amplifier (LNA) in the receivepath or a power amplifier (PA) in the transmit path. Another commonembodiment for FDD has certain directive gain antenna elements 352 usedonly for transmit and then Front-end 408 for such transmit antennaelements would have a PA and one or more RF filters for a transmit FDDsub-band and has certain directive gain antenna elements 352 used onlyfor receive and then Front-end 408 for such receive antenna elementswould have an LNA and one or more RF filters for a receive FDD sub-band.In most ZDD embodiments, certain directive gain antenna elements 352 areused only for transmit and others only for receive with respectiveFront-ends as described for FDD except that the RF filters overlap inthe frequency domain for both transmit and receive (i.e. no separatetransmit and receive sub-bands).

Note that each antenna 352 has a directivity gain Gq. For IBRs intendedfor fixed location street-level deployment with obstructed LOS betweenIBRs, whether in PTP or PMP configurations, each directive gain antenna352 may use only moderate directivity compared to antennas inconventional PTP systems at a comparable RF transmission frequency. Asdescribed in greater detail in U.S. patent application Ser. No.13/645,472, U.S. patent application Ser. No. 13/898,429, and U.S. patentapplication Ser. No. 14/108,200, typical values of Gq are on the orderof 10 to 20 dBi for each antenna at RF transmission frequencies below 10GHz.

In the IBR Antenna Array 348, the total number of individual antennaelements 352, Q, is at least greater than or equal to the larger of thenumber of RF transmit chains 336, M, and the number of RF receive chains340, N. In some embodiments, some or all of the antennas 352 may besplit into pairs of polarization diverse antenna elements realized byeither two separate feeds to a nominally single radiating element or bya pair of separate orthogonally oriented radiating elements. In someembodiments, certain antenna elements 352 may be configured withdifferent antenna gain Gq and/or radiation patterns compared to othersin the same IBR. Also, in many embodiments, such as for those employingFDD or ZDD, U.S. patent application Ser. No. 13/645,472, U.S. patentapplication Ser. No. 13/898,429, and U.S. patent application Ser. No.14/108,200 provide additional description of advantageous arrangementsof separate transmit and receive antenna subsets with the total set Q ofindividual antenna elements 352.

The IBR RF Switch Fabric 412 provides selectable RF connections betweencertain RF-Tx-m and/or certain RF-Rx-n to the various individual antennaelements 352 via various front-end 408 embodiments. Note specificallythat in certain embodiments the individual antenna elements 352 arecoupled via a transmit-only front-end and/or the IBR RF Switch Fabric412 to only a transmit chain output RF-Tx-m or coupled via areceive-only front-end and/or the IBR RF Switch Fabric 412 to only areceive chain output RF-Rx-n to advantageously enable separateoptimization of the receive antenna array from that of the transmitantenna array. U.S. patent application Ser. No. 13/645,472, U.S. patentapplication Ser. No. 13/898,429, and U.S. patent application Ser. No.14/108,200 provide additional description of different embodiments ofthe IBR RF Switch Fabric 412 as applicable to TDD, FDD and ZDD indifferent product configurations.

With reference back to FIG. 3, the IBR RF 332 also includes transmit andreceive chains 336, 340. In one embodiment, each element of transmitchain 336 takes a transmit chain input signal such as digital basebandquadrature signals I_(Tm) and Q_(Tm) and then converts them to atransmit RF signal RF-Tx-m at an RF carrier frequency typically below 10GHz. Similarly, each element of receive chain 340 converts a receive RFsignal RF-Rx-n at an RF carrier frequency typically below 10 GHz to areceive chain output signal such as digital baseband quadrature signalsI_(Rn) and Q_(Rn).

Other IBR elements include the IBR MAC 312, the Radio Link Control (RLC)356, the Radio Resource Control (RRC) 360 and the optional IBMS Agent.Although IBR embodiments are possible wherein the MAC 312, RLC 356, RRC360 and the optional IBMS Agent are distinct structural entities, morecommonly IBRs are realized wherein the MAC 312, RLC 356, RRC 360 and theoptional IBMS Agent as well as portions of the IBR Interface Bridge 308are software modules executing on one or more microprocessors. Note alsothat in some IBR embodiments that use of a “Software Defined Radio”(SDR) for the IBR Modem 324 and/or IBR Channel MUX 328 or portionsthereof may also be realized in software executing on one or moremicroprocessors. Typically in SDR embodiments, the one or moremicroprocessors used for elements of the PHY layer are physicallyseparate from those used for the MAC 312 or other layers and arephysically connected or connectable to certain hardware cores such asFFTs, Viterbi decoders, DFEs, etc. As SDR processing power increasesover time, functions traditionally implemented in hardware coresadvantageously migrate to the SDR processor cores as software modulesfor greater implementation flexibility.

The RRC 360 and RLC 356 interact with the IBR MAC 312 and variouselements of the IBR PHY both via “normal” frame transfers and directcontrol signals via the conceptual IBR Control plane. Both the RRC 360and the RLC 356 may execute concurrent control loops with the respectivegoals of optimizing radio resource allocations and optimizing radio linkparameters for current resources in view of the dynamic propagationenvironment conditions (including uncoordinated interference ifapplicable), IBR loading, and possibly system-wide performance goals(via the optional IBMS Agent or other IBR to IBR control communicationslinks). It is instructive to view the RLC 356 as an “inner loop”optimizing performance to current policies and radio resourceallocations for each active link and to view the RRC 360 as an “outerloop” determining if different policies or radio resource allocationsare desirable to meet overall performance goals for all IBRs currentlyinteracting with each other (intentionally or otherwise). Typically boththe RRC 360 and the RLC 356 are implemented as software modulesexecuting on one or more processors.

The primary responsibility of the RLC 356 in exemplary IBRs is to set orcause to be set the current transmit Modulation and Coding Scheme (MCS)and output power for each active link. The RLC 356 causes the transmitpower control (TPC) of the IBR to be maintained both in a relative senseamongst active links, particularly of interest for the AE-IBR in a PMPconfiguration, and also in an overall sense across all transmits chainsand antennas.

In some embodiments, the RLC 356 can determine its MCS and TPCselections across active links based on information from various sourceswithin the IBR. For example, the IBR MAC can deliver RLC control framesfrom other IBRs with information from such other IBRs (for example,RSSI, decoder metrics, FCS failure rates, etc.) that is useful insetting MCS and TPC at the transmitting IBR. Additionally, such RLCcontrol frames from an associated IBR may directly request or demandthat the RLC in the instant IBR change its MCS and/or TPC values fortransmit directly on either a relative or absolute basis. U.S. patentapplication Ser. No. 13/645,472 and U.S. patent application Ser. No.14/108,200 provide additional description of different embodiments ofthe RLC 356 as applicable to TDD, FDD and ZDD in different productconfigurations.

The primary responsibility of the RRC 360 is to set or cause to be setat least the one or more active RF carrier frequencies, the one or moreactive channel bandwidths, the choice of transmit and receive channelequalization and multiplexing strategies, the configuration andassignment of one or more modulated streams amongst one of moremodulator cores, the number of active transmit and receive RF chains,and the selection of certain antenna elements and their mappings to thevarious RF chains. Optionally, the RRC may also set or cause to be setthe superframe timing, the cyclic prefix length, and/or the criteria bywhich blocks of Training Pilots are inserted. The RRC 360 allocatesportions of the IBR operational resources, including time multiplexingof currently selected resources, to the task of testing certain linksbetween an AE-IBR and one or more RE-IBRs. The RRC 360 evaluates suchtests by monitoring at least the same link quality metrics as used bythe RLC 656. Additionally, in some embodiments, additional RRC-specificlink testing metrics are also used. The RRC 360 can also exchangecontrol frames with a peer RRC at the other end of an instant link to,for example, provide certain link testing metrics or request or directthe peer RRC to obtain link specific testing metrics at the other end ofthe instant link for communication back to RRC 360.

In some embodiments, the RRC 360 causes changes to current resourceassignments in response to tested alternatives based on policies thatare configured in the IBR and/or set by the optional IBMS Agent. Anexemplary policy includes selecting resources based on link qualitymetrics predicted to allow the highest throughput MCS settings at lowestTPC value. Additional exemplary policies may factor in minimizinginterference by the instant link to other AE-IBR to RE-IBR links (orother radio channel users such as conventional PTP radios) eitherdetected at the instant IBRs or known to exist at certain physicallocations nearby as set in configuration tables or communicated by theoptional IBMS Agent or other IBR to IBR control communications links asdescribed, for example, in co-pending U.S. patent application Ser. No.14/098,456, the entirety of which is hereby incorporated by reference.For example, U.S. patent application Ser. No. 14/098,456 disclosesexemplary systems and methods for control communications links in theform of inline or embedded signals that may be suitable for exchange ofcontrol information between IBRs that otherwise lack any IBR to IBRcommunication path. Such policies may also be weighted proportionatelyto reach a blended optimum choice amongst policy goals or rankedsequentially in importance.

In some embodiments, for either PTP or PMP deployment configurations,the selection of either the one or more active RF carrier frequenciesused by the RF chains of the IBR RF, the one or more active channelbandwidths used by the IBR MAC, IBR Modem, IBR Channel MUX and IBR RF,the superframe timing, the cyclic prefix length, or the insertion policyfor blocks of Training Pilots is determined at the AE-IBR for any givenlink. The RE-IBR in such an arrangement can request, for example, an RFcarrier frequency or channel bandwidth change by the AE-IBR by sendingan RRC control frame in response to current link conditions at theRE-IBR and its current RRC policies. Whether in response to such arequest from the RE-IBR or due to its own view of current linkconditions and its own RRC policies, an AE-IBR sends the affectedRE-IBRs an RRC control frame specifying at least the parameters for thenew RF frequency and/or channel bandwidth of the affected links as wellas a proposed time, such as a certain superframe sequence index, atwhich the change-over will occur (or alternatively, denies the request).The AE-IBR then makes the specified change after receiving confirmationRRC control frames from the affected RE-IBRs or sends a cancellation RRCcontrol frame if such confirmations are not received before thescheduled change.

An RE-IBR typically attempts to utilize all available modulator anddemodulator cores and streams as well as all available RF chains tomaximize the robustness of its link to a particular AE-IBR. In an RE-IBRembodiment where at least some redundancy in antenna elements amongstspace, directionality, orientation, polarization and/or RF chain mappingis desirable, the primary local RRC decision is then to choose amongstthese various antenna options. In other embodiments the AE-IBR andRE-IBR optimize their resource allocations independently such that thereis little distinction between the RRC strategies at the AE-IBR versusthe RE-IBR. U.S. patent application Ser. No. 13/645,472, U.S. patentapplication Ser. No. 13/898,429, and U.S. patent application Ser. No.14/108,200 provide additional description of different embodiments ofthe RRC 360 as applicable to TDD, FDD and ZDD in different productconfigurations.

The specific details of the IBR Modem 324 and IBR Channel MUX 328 dependsomewhat on the specific modulation format(s) deployed by the IBR. Ingeneral, the IBR requires a modulation format suitable for a broadbandchannel subject to frequency-selective fading and multipathself-interference due to the desired PHY data rates and ranges inobstructed LOS propagation environments. Many known modulation formatsfor such broadband channels are possible for the IBR. Two suchmodulation formats for the IBR are (1) Orthogonal Frequency DivisionMultiplexing (OFDM) and (2) Single-Carrier Frequency Domain Equalization(SC-FDE). Both modulation formats are well known, share commonimplementation elements, and have various advantages and disadvantagesrelative to each other. U.S. patent application Ser. No. 13/645,472provides additional detail regarding OFDM and SC-FDE as applicable tovarious IBR embodiments.

The specific details of the IBR Modem 324 and IBR Channel MUX 328 alsodepend somewhat on the specific antenna array signal processingformat(s) deployed by the IBR. In general, the IBR utilizes multipleantennas and transmit and/or receive chains, which can be utilizedadvantageously by several well-known baseband signal processingtechniques that exploit multipath broadband channel propagation. Suchtechniques include Multiple-Input, Multiple-Output (MIMO), MIMO SpatialMultiplexing (MIMO-SM), beamforming (BF), maximal ratio combining (MRC),and Space Division Multiple Access (SDMA). U.S. patent application Ser.No. 13/645,472 provides additional detail regarding such techniques asapplicable to various IBR embodiments.

In many embodiments, the IBR Modem 324 comprises one or modulator coreseach of which comprises such functional elements as scramblers,encoders, interleavers, stream parsers, symbol groupers and symbolmappers. At a high level, each modulator core within the IBR Modem 324typically transforms a data stream from the IBR MAC 312 into a symbolstream that can be passed to the IBR Channel MUX 328. Similarly, in manyembodiments, the IBR Modem 324 also comprises one or demodulator coreseach of which comprises such functional elements as descramblers,decoders, deinterleavers, stream multiplexers, and soft decision symboldemappers. At a high level, each demodulator core within the IBR Modem324 typically transforms a stream of estimated receive symbols, such asrepresented by a Log-Likelihood Ratio (LLR), from the IBR Channel MUX328 into a data stream that can be passed to the IBR MAC 312. U.S.patent application Ser. No. 13/645,472, U.S. patent application Ser. No.13/898,429, and U.S. patent application Ser. No. 14/108,200 provideadditional description of different embodiments of the IBR Modem 324 asapplicable to TDD, FDD and ZDD in different product configurations.

In many embodiments, the IBR Channel MUX 328 comprises a transmit pathchannel multiplexer that may or may not be frequency selective and thatin turn may comprise such functional elements as block assemblers,transmit channel equalizers, transmit multiplexers, cyclic prefixadders, block serializers, transmit digital front ends, preambleinserters, and pilot inserters. At a high level, the transmit path ofthe IBR Channel MUX 328 transforms one or more symbol streams from theIBR Modem 324 into inputs for the one or more transmit chains eachcomprised of baseband symbol samples. Similarly, in many embodiments,the IBR Channel MUX 328 also comprises a frequency selective receivepath channel multiplexer that in turn may comprise that in turncomprises such functional elements as synchronizers, receive digitalfront ends, cyclic prefix removers, channel equalizer coefficientsgenerators, receive channel equalizers, receive stream multiplexers andcomplex Discrete Fourier Transformers (DFT). At a high level, thereceive path of the IBR Channel MUX 328 transforms the outputs of theone or more receive chains each comprised of baseband symbol samplesinto one or more streams of estimated receive symbols for input into theIBR Modem 324. U.S. patent application Ser. No. 13/645,472, U.S. patentapplication Ser. No. 13/898,429, and U.S. patent application Ser. No.14/108,200 provide additional description of different embodiments ofthe IBR Channel MUX 328 as applicable to TDD, FDD and ZDD in differentproduct configurations.

In exemplary embodiments, the IBR MAC 312 comprises such functionalelements as a management entity, a Tx buffer and scheduler, a controlentity, an Rx buffer, a frame check sum (FCS) generator, a headergenerator, a header analyzer and an FCS analyzer. U.S. patentapplication Ser. No. 13/645,472, U.S. patent application Ser. No.13/898,429, and U.S. patent application Ser. No. 14/108,200 provideadditional description of different embodiments of the IBR MAC 312 asapplicable to TDD, FDD and ZDD in different product configurations.

Additional details regarding numerous optional functional components andregarding additional exemplary embodiments of the IBR are provided incommonly assigned U.S. patent application Ser. No. 13/645,472, U.S. Pat.No. 8,311,023 and U.S. Pat. No. 8,238,318, U.S. patent application Ser.No. 13/898,429 and U.S. Pat. No. 8,467,363, U.S. patent application Ser.No. 13/271,051 and U.S. Pat. No. 8,300,590, and U.S. patent applicationSer. No. 14/108,200 and U.S. Pat. Nos. 8,638,839 and 8,422,540, thedisclosures of which are hereby incorporated herein by reference intheir entirety.

Antenna assembles having improved feed mechanisms to address theseproblems will now be described. The patch array antenna assemblyincludes an array of resonant radiating patch antenna elements that areaperture-fed (instead of pin or probe-fed). In this improved antennadesign, the feed network is coupled to the resonant radiating cavity viaapertures in the conductive ground plane. In one embodiment, the feednetwork is composed of transmission lines such as, for example,microstrip lines on one side of a printed circuit board (PCB) whereinthe conductive ground plane is on the other side of the PCB. In anotherembodiment, the feed network is composed of transmission lines such as,for example, striplines within a multi-layer PCB wherein at least onelayer is a conductive ground plane that includes the apertures on theoutside of the PCB. Exemplary advantages of an aperture-fed antennaarray include, for example, lower cost due to reduced assemblytime/complexity and higher reliability due to no solder joints securingany pins. The shape of the aperture also provides distinct advantagesover the prior art. In particular, by using an aperture having amodified shape (e.g., rounded or tapered) compared to a conventional art“H” or “dogbone” slot shape, the wanted coupling to the resonantradiating cavity is maximized and unwanted backwards facing radiationfrom is minimized. The use of a multi-layer PCB and a striplinetransmission line feed network further minimizes the unwanted backwardsfacing radiation as well as enables greater flexibility for placingactive electronic components on the opposite side of the PCB from theside with the apertures in the conductive ground plane.

Another improvement is the termination of the feed line. In conventionalaperture-fed patch antennas the magnetic (inductive) coupling betweenthe transmission line feed and resonant radiating cavity results inexcessive inductive reactance at the antenna feed. To counter thisinductive reactance, the feed line is commonly left open-stub aftercrossing the aperture to provide a series capacitive reactance. Thisopen-stub tuning has the undesirable side effect of increasingbackwards-facing radiation. By co-optimizing the conductive patchelement dimensions and aperture dimensions, the feed can be tuned forzero-net reactance, and a desired input resistance (e.g. 50 ohms, or 100ohms). This allows the feed line to be terminated in a short circuitimmediately after crossing the aperture, resulting in lowerbackwards-facing radiation than the conventional art open circuitcounterpart.

In some embodiments, a dielectric spacer is provided between the twoPCBs to provide the desired spacing between the conductive patchelements and conductive ground plane thereby forming a resonantradiating cavity. In an exemplary embodiment, this spacer can be asimple injection-molded plastic part. The spacer includes one or moresymmetric openings that remove any dielectric material from within theresonant radiating cavity between the conductive patch element andconductive ground plane. Dielectric material within the resonantradiating cavity is undesirable because any variation in the materialdielectric constant can cause the resonant frequency of the resonantradiating cavity to shift, thereby reducing radiation efficiency.Additional details of this patch array antenna assembly design will nowbe discussed with reference to several Figures. The exemplary antennaassembly embodiments described herein may be used in the IBR embodimentsdescribed above and in the incorporated co-pending applications as apair of directive gain antenna elements for a facet comprising a firstdirective gain antenna element with a first polarization and a seconddirective gain antenna element with a second polarization that isorthogonal to the first polarization.

FIG. 5A illustrates a patch array antenna assembly 500 in accordancewith some embodiments of the invention. A detailed view of the side ofthe antenna assembly is shown in FIG. 5B.

As shown in FIG. 5A, the patch array antenna assembly 500 includes afirst substrate 504, a spacer 508 and a second substrate 512. The firstsubstrate 504, spacer 508 and second substrate 512 have approximatelythe same overall width and length and are approximately aligned with oneanother.

In FIG. 5A, a plurality of rivets 516 are inserted into openings 520 inthe first substrate 504, spacer 508 and second substrate 512 to hold thefirst substrate 504, spacer 508 and second substrate 512 together. Itwill be appreciated that other methods may be used to secure or hold thefirst substrate 504, spacer 508 and second substrate 512 together. Forexample, rivets having an alternative shape, different fasteners (e.g.,screws, bolts, clamps, etc.), adhesives or other methods known to thoseof skill in the art may be used to hold the first substrate 504, spacer508 and second substrate 512 together. It will further be appreciatedthat fewer than or more than the number of rivets 516 shown in FIG. 5Amay be used to secure or hold the first substrate 504, spacer 508 andsecond substrate 512 together.

The first substrate 504 is typically a printed circuit board (PCB). Insome embodiments, the first substrate 504 is at least a conductive patchelement carrier. A number of conductive patch elements may be located onone side of the first substrate 504. In some embodiments, the pluralityof conductive patch elements is located on the surface of the firstsubstrate 504 adjacent to the spacer 508. The conductive patch elements,as will be described in further detail hereinafter, define a number ofresonant radiating patch antenna elements.

The second substrate 512 is also typically a printed circuit board(PCB). The second substrate 512 includes a transmission line feednetwork for the patch array antenna assembly. The top outer layer of thesecond substrate 512 may also be a conductive ground plane 528 that, incombination with the conductive patch elements of the first substrate,forms resonant radiating patch antenna elements. As will be described infurther detail herein, the conductive ground plane 528 on the secondsubstrate 512 includes a plurality of apertures (or openings) forcoupling the resonant radiating cavities to the transmission line feednetwork on the second substrate 512.

The spacer 508 is positioned between the first substrate 504 and thesecond substrate 512. The spacer 508 has a thickness that is selected tomaintain the proper height within the resonant radiating cavity (saidheight being the distance between the conductive ground plane 528 on thesecond substrate 512 and the conductive patch elements on the firstsubstrate 504 in some embodiments) as shown in FIG. 5A. Alternatively,in other embodiments where the conductive patch elements on the firstsubstrate 504 are on the opposite side of substrate 504 from the surfaceadjacent to spacer 508 or on an inner layer of a multi-layer PCB, theheight within the resonant radiating cavity is equal to the thickness ofthe spacer 508 plus the thickness of any PCB layers between the spacerand the surface with the conductive patch elements.

In FIG. 5A, the spacer 508 includes two openings 524A, 524B wheredielectric material is absent. The spacer 508 and the openings 524, inparticular, ensure there is no dielectric material in the resonantradiating cavity, other than air, making the structure more robustagainst variations in the electrical properties of the spacer material.Although two openings 524A, 524B are shown in FIG. 5A, it will beappreciated that the spacer 508 may include fewer than two or more thantwo openings 524. For example, the spacer 508 may include one opening.In another example, the spacer 508 may include three or more openings.In another example, the number of openings in the spacer 508 may beequal to the number of conductive patch elements on the first substrate504. Alternatively, the number of openings in the spacer 508 may be onequarter or one half the number or any other number of conductive pathelements on the first substrate 504. Further, the shape of the openingsmay differ from that shown in FIG. 5A. For example, if one opening isprovided for each conductive patch element of circular shape, then thespacer openings may also be circular but larger in diameter than that ofthe circular conductive patch element. In another example, if the shapeof the conductive patch element is non-circular and there is aone-to-one correspondence of openings to conductive patch elements, theshape of the opening may be the same shape as the conductive patchelement but also a larger dimension than the conductive patch element.

In order to minimize the effect of the spacer electrical properties onthe antenna performance, the spacer opening should be larger than theconductive patch element (or alternatively, the spacer opening shouldextend beyond the projected area of the conductive patch element) by atleast the spacer thickness and preferably the spacer opening should belarger than the conductive patch element by two times the spacerthickness. Additional differences in shape and size will be understoodby one of skill in the art and additional details regarding the spacer508 will be described hereinafter. This approach allows more flexibilityin the choice of spacer material than the conventional-art (i.e., asolid spacer) because some variation in the dielectric parameters of thespacer 508 can be tolerated with minimal effect on the patch arrayantenna assembly. This also allows the use of less expensive materialsfor fabrication of the spacer 508 than the conventional-art (i.e., asolid spacer).

FIG. 5C shows an exemplary patch antenna assembly that shares the sameconstruction features as shown in FIGS. 5A and 5B, but with the additionof a second instance of spacer 508 and a third substrate 532. Thisassembly interleaves two spacers, 508A and 508B, between a firstsubstrate 504, a third substrate 532, and a second substrate 512. Thirdsubstrate 532 is a conductive patch element carrier, having a pluralityof conductive patch elements located on either or both of its twosurfaces. In some embodiments, the number of conductive patch elementson the third substrate 532 is equal to the number of conductive patchelements on the first substrate 504, and the plurality of conductivepatch elements on the third substrate 532 are concentric with therespective ones of the plurality of conductive patch elements on thefirst substrate 504 (or alternatively, coincident with a projection ofthe plurality of conductive patch elements on the first substrate 504).In some embodiments, the conductive patch elements on the firstsubstrate 504 are of a larger area (or diameter if circular) than theconductive patch elements on the third substrate 532. In an embodimentwith circular conductive patch elements, the diameter of the conductivepatch elements on the first substrate 504 is typically 10% larger thanthe diameter of the conductive patch elements on the third substrate532.

The conductive patch elements on the first substrate 504, the conductivepatch elements on the third substrate 532, and conductive ground plane528 on the second substrate 512 form stacked resonant radiating patchantenna elements. A stacked resonant radiating patch antenna elementprovides wider bandwidth than can be achieved with radiating patchantenna elements comprising a single conductive patch element.Typically, a conventional, single conductive patch element can achieve aresonant radiating patch antenna element with an impedance bandwidth ofabout 5% of the target operating frequency, whereas a stacked resonantradiating patch antenna element, as described herein, can achieve animpedance bandwidth of up to 20% of the target operating frequency.

In some embodiments, one or both of first substrate 504 and thirdsubstrate 532 are formed of a dielectric film material, such as, forexample, polyimide. It will be appreciated that other dielectric filmmaterials may used to form the first substrate 504 and/or thirdsubstrate 532. The conductive patch elements may be formed by a copperdeposition process, such as, for example, copper sputtering. It will beappreciated that alternative methods may be used to form the conductivepatch elements. Dielectric film materials are advantageous because theycan be used to form a very thin substrate (e.g., 0.1 mm), which, inturn, minimizes undesirable dielectric loading of the resonant radiatingcavities by the substrate material. In other embodiments, one or both ofthe first substrate 504 and third substrate 532 are formed from aprinted circuit board. In yet other embodiments, one of either firstsubstrate 504 or third substrate 532 is formed from a printed circuitboard while the other is formed from a dielectric film material. Theexemplary aperture feeding techniques disclosed below are compatiblewith this combination three-substrate and two-spacer assembly. In someembodiments, the conductive patch elements 604 are etched in the firstsubstrate 504 using known techniques and known materials. In oneembodiment, the conductive patch elements 604 are formed from a metalsuch as etched copper elements.

FIG. 6 shows a bottom view of the first substrate 504. As shown in FIG.6, the first substrate 504 includes a surface 600 on which conductivepatch elements 604 are located. When assembled, the conductive patchelements 604 on the first substrate 504 and the conductive ground plane528 on the second substrate 512 form resonant radiating patch antennaelements that can be excited by aperture feeds as described herein.These resonant radiating patch antenna elements support simultaneousexcitation in multiple, orthogonal, electromagnetic modes that cancorrespond to vertical polarization and horizontal polarization ordual-slant 45-degree polarization for the respective far-field directivegain antenna patterns. With the aperture design as described herein, thecoupling between the orthogonal modes can be very low.

In some embodiments, the combination of the feed network and theplurality of resonant radiating patch antenna elements form a phasedarray. In a phased array embodiment, the resonant radiating patchantenna elements are excited in a specific relative phase and amplitudeto attain different performance characteristics than what is realizablefrom a single resonant radiating patch antenna element. Exemplaryperformance characteristics of a phased array antenna assembly arehigher far-field pattern gain, improved spatial selectivity, andincreased control over coupling to nearby antennas as also described ingreater detail in U.S. patent application Ser. No. 13/898,429 and U.S.Pat. No. 8,467,363, the disclosures of which are hereby incorporatedherein by reference in their entirety.

In the embodiment depicted in FIG. 6, the conductive patch elements 604are arranged in an array of four rows with two columns where eachrow/column combination corresponds to one conductive patch element 604within an array. In the embodiment of FIG. 6, the four rows ofconductive patch elements 604 in the array are arranged such that theaperture feeds described herein can provide two orthogonal directivegain antenna elements of a patch array antenna assembly each withincreased directive gain in the elevation pattern compared to an antennaformed by a single row of one or more resonant radiating patch antennaelements. Similarly, the two columns of conductive patch elements 604 inthe array are arranged such that the aperture feeds described herein canprovide two orthogonal directive gain antenna elements of an antennaassembly each with increased directive gain in the azimuthal patterncompared to an antenna formed by a single column of one or more resonantradiating patch antenna elements. Other array embodiments (not shown)may have only a single column of conductive patch elements 604 or mayhave more than two columns of conductive patch elements 604 or may haveeither more or less than four rows of conductive patch elements 604. Itwill be further appreciated that the first substrate 504 may includefewer than eight or more than eight conductive patch elements 604.

In the embodiment depicted in FIG. 6, the conductive patch elements 604are located on the bottom surface (that is, the surface adjacent to thespacer when assembled) of the first substrate 504 so that, whenassembled, the material that forms the spacer 508 is not within theresonant radiating cavity formed by the conductive patch elements 604and the conductive ground plane 528. Those skilled in the art willrecognize that the conductive patch elements 604 can alternately belocated on the top surface of the first substrate 504. This arrangementis less desirable, since the dielectric parameters of the firstsubstrate 504 will more heavily influence the resonant frequency of theresonant radiating cavity, or hence each resonant radiating patchantenna element. This dielectric loading can be accounted for in designof the conductive patch element shape and size, but in practice,variability in the dielectric parameters of the first substrate willthen cause undesirable variability in resonant frequency of the resonantradiating patch antenna element.

FIG. 7 illustrates an exemplary arrangement of the spacer 508 relativeto the bottom surface of the first substrate 504. As shown in FIG. 7,the openings 524 in the spacer 508 are designed to coordinate with theconductive patch elements 604. In particular, the opening 524A isaligned with four of the patch elements 604, and opening 524B is alignedwith the other four patch elements 604 on the bottom surface of thefirst substrate 504. As explained above, each of the openings 524A, 524Bis designed to ensure the spacer does not significantly impinge into theresonant radiating cavity formed between the conductive patch elementand the conductive ground plane. For example, in a specific embodimentoptimized for operation at 5.3 GHz, the patch elements have a radius of12.42 mm, the spacer thickness is 2.4 mm and the spacer opening islarger than the projected area of each conductive patch element by atleast 5.08 mm.

As shown in FIG. 7, the openings are shown having a rectangular shapehaving rounded edges. However, it will be appreciated that alternativeshapes may be used. For example, the openings may be circular, theradius of the curve may be less than or greater than that shown in FIG.7, the openings may be rectangular, etc., as understood by those ofskill in the art.

In one embodiment, the openings 524 in the spacer 508 are designed formultiple antenna assemblies that operate at differing target operatingfrequencies. As understood by those of skill in the art, the dimensionsand relative position of the conductive patch elements differ dependingon the desired target operating frequency of the antenna assembly. Bycorrectly sizing the openings in the spacer, one spacer may be used withantenna assemblies for these differing operating frequencies. Forexample, the same spacer 508 may be used with a substrate 504 that isconfigured for a 5.3 GHz target operating frequency, a 5.6 GHz targetoperating frequency and a 5.8 GHz target operating frequency. In thisexample, setting the spacer opening dimensions sufficiently large enoughto cause negligible antenna performance variation effects to the 5.3 GHzconductive patch element, as described above, also causes the spaceropening to be large enough for the 5.6 GHz and 5.8 GHz optimizeddesigns.

FIG. 8A is a view of the second substrate 512 illustrating both a firstlayer or “top surface” that comprises a plurality of pairs of apertures820, 824 being repeated across the conductive ground plane 528 and asecond layer or “bottom surface” that comprises transmission line feednetworks such as 804A and 804B. As shown in FIG. 8A, the secondsubstrate 512 includes microstrip transmission line feed networks 804Aand 804B each with respective feed points 836A and 836B. As understoodby those of skill in the art, these feed networks might also be realizedas stripline transmission line structures if the second substrate were amulti-layered printed circuit board with more than two layers andmultiple ground planes. In FIG. 8A, radio transceiver electronicscomponent placement patterns 808 a and 808 b are co-located with themicrostrip feed networks 804 on the bottom side of the printed circuitboard substrate 512 which in this embodiment corresponds to the secondlayer. The co-location of radio transceiver components on the samesubstrate as the feed network provides a very short, and in turnlow-loss, interconnect between the feed points 836A and 836B and therest of the radio components. For example, co-locating a low noiseamplifier and preferably also at least one RF bandpass filter on thesame substrate as the feed network is advantageous because thisincreases the amount of loss in cables that may be part of the couplingbetween directive gain antenna elements and receive RF chains that canbe tolerated without any degradation in the radio link performance.

In some embodiments, electronic components may be located within thefeed networks 804. The integration of electronic components for tunableelements such as tunable capacitors in series or shunt with lumpedelement or distributed circuit elements within the feed network allowsdynamic adjustment of the characteristics of the patch array antennaassembly, such as far-field radiation patterns, cross-coupling betweenthe orthogonal polarizations as measured at their feed points, orcoupling to nearby antennas, in order to optimize a desired radio linkmetric, such as a signal to noise and/or interference ratio or such as adegree of isolation between an directive gain antenna element used fortransmit and another used for receive under full duplex operationconditions. In an exemplary IBR embodiment, the RRC 360 (or some othercontroller such as a ZDD Canceller Loop Coefficients Generator describedin U.S. patent application Ser. No. 14/108,200 and U.S. Pat. Nos.8,638,839 and 8,422,540, the disclosures of which are herebyincorporated herein by reference in their entirety) provides or causesto be provided an input signal to the tunable element so that theantenna assembly characteristic is adjusted according to the desiredmetric.

FIG. 8A shows a plurality of pairs of apertures 820, 824 being repeatedacross the conductive ground plane 528 on the top surface, or firstlayer, of the second substrate in a pattern that matches thedistribution of conductive patch elements on the first substrate, asshown within an array in FIG. 7. Preferably, the apertures are dividedinto pairs 820, 824. Each pair of apertures includes a first aperture820 in a first direction, and a second aperture 824 in a seconddirection that is orthogonal to the first aperture 820. In someembodiments, each pair of apertures 820, 824 corresponds to one of theconductive patch elements 604 on the first substrate 504. Each aperture820 or 824 excites its corresponding resonant radiating patch antennaelement in a single respective electromagnetic mode, and the respectivemodes are then orthogonal to each other in an electromagnetic sense. Thetwo apertures in each pair of apertures are arranged such that the endpoint of a first aperture 824 is aligned with the mid-point of a secondaperture 820, in a T-shape. This T-shape arrangement achieves a very lowcoupling between the two orthogonal apertures as compared with the morecommon L-shape arrangement of the conventional art where the twoapertures are aligned at one endpoint. Such desirable lower couplingoccurs at least because the relative excitation of the T-shapearrangement is in common mode which causes a cancellation effect to theother aperture.

As shown in FIG. 8A, the left hand column of vertical apertures 824 isfed from the right-hand side. This means that each respective feedlineportions 812A to 812D overlap each aperture by crossing over arectangular aperture body for each respective left hand verticalaperture 824 from a center point on the right of the rectangularaperture body to a termination point 828 on the left of the rectangularaperture body, as also illustrated in FIG. 8B. Similarly, the right handcolumn of vertical apertures 824 is fed from the left-hand side. Thus,in some embodiments, the vertical apertures depicted in FIG. 8A forexcitation of the resonant radiating patch antenna elements set by thearray of conductive patch elements depicted in FIG. 6 are fed in anopposite direction for the first or left hand column of respective pairsof apertures and conductive patch elements from the direction for thesecond or right hand column of respective pairs of apertures andconductive patch elements. The feedline portions 812A through 812D mustbe offset by an electrical length of 90 degrees towards one column tophase align the electrical modes excited in the two columns ofconductive patch elements in order to achieve the desired arrayproperties. The feedline portions 812A to 812D are shown with an offsetof 90 degrees equivalent electrical length at the target operatingfrequency to the left-hand side of the “center” of each of the feedlineportions 812A to 812D, but could also be offset by 90 degrees to theright hand side. This feed arrangement is advantageous because itminimizes the space needed on the substrate 512 so that more feeds,apertures and conductive patch element combinations can be included inthe patch array antenna assembly, thereby increasing the potential arraygain that can be realized from a given substrate size.

As shown in FIG. 8A, one or more of the feed lines may also include ameandering line portion 832. The meandering line portion 832 may includeone or more bends so that the physical distance or electrical length(and hence the group delay) is the same from each common feed point 836Aor 836B to each respective aperture 824 or 820 in the array structureConventional art approaches, such as series feeding techniques, matchonly the relative phase of each aperture—not the group delay, like thefeeding technique illustrated in FIG. 8A. The matched group delayapproach (e.g., using the feed line having a meandering line portionshown in FIG. 8A) is advantageous because the resulting feed networkmaintains proper phase excitation of the array of resonant radiatingpatch antenna elements independently of frequency, thereby resulting ina very broadband feed structure.

Additionally, as shown in FIG. 8A, the feed lines 804 are terminated ina short circuit. By tuning the aperture width as described herein, thefeed lines 804A and 804B can be terminated in a short circuit—ratherthan an open stub as in the conventional art—using the vias 828. Thevias 828 extend from the feed network 804A and 804B on one surface ofthe second substrate 512 (or optionally, if stripline, from a feednetwork on an inner layer) and through to the ground plane to theopposing surface of the second substrate 512 to form a short circuittermination, thereby realizing the advantage of lower backwards facingradiation as described above.

FIG. 8B further illustrates the offset arrangement of the feed network.As shown in FIG. 8B, horizontal apertures 820 are fed by a feedlineportion 852, which splits into two feedline portions 856 and 860. InFIG. 8B, feedline portions 856 and 860 are the same distance from the3-way T-junction joining feedline portions 852, 856 and 860, and theresulting symmetric structure ensures horizontal apertures 820 areexcited by feedline portions that overlap the rectangular aperture bodyby overlapping (via crossing over) the rectangular aperture body in thesame direction with the same relative phase angle as is desirable forthe array performance in this exemplary embodiment. As shown in FIG. 8B,vertical apertures 824 are fed by a feedline portion 848 which splitsinto feedline portions 840, 844. In FIG. 8B, the feedline portion 844 iselectrically longer than the feedline portion 840 by a distanceequivalent to 180 degrees in phase to correct for the phase offset ofthe mirrored feeds (left-hand versus right-hand) as described above.

FIG. 8B further illustrates the short circuit termination. As shown inFIG. 8B, each of the feedline portions 856, 860, 840, 844 terminateswith a via 828 that connects to the conductive ground plane 528. Asexplained above, the vias 828 result in short circuit terminations ofeach of the feedline portions.

In some embodiments, as shown in FIG. 8C, a first layer of the secondsubstrate 512 is located at the surface adjacent to spacer. In FIG. 8C,the second substrate 512 includes a ground plane with a plurality ofapertures 820, 824. Other openings in the conductive ground plane 528are locations of drilled holes used for assembly purposes and are notadditional apertures within the antenna assembly.

In some embodiments, as shown in FIG. 8D, a second layer is located atthe surface opposite to spacer 508 of the second substrate 512,including first and second transmission line feed networks 804A and804B. In other embodiments that use a multi-layer PCB, the portion ofFIG. 8D showing the first and second transmission line feed networks804A and 804B may be located on an inner layer between two ground planesusing stripline structures instead of microstrip.

FIG. 8E is a detailed view of the apertures 820, 824 that are used tofeed the resonant radiating patch antenna elements. As shown in FIG. 8E,the aperture is defined by a rectangular aperture body 864 and twoaperture ends 868A and 868B. The aperture ends 868A and 868B are widerthan the width of the rectangular aperture body 864. In someembodiments, as shown in FIG. 8E, the edges of the aperture ends 868 aretapered or rounded.

The rectangular aperture body 864 has an aperture body width w that ischosen to be as narrow as can be reliably fabricated by standard printedcircuit board construction capabilities for the selected PCB that formsthe second substrate 512. Under current processing capabilities, anaperture body width w of about as low as 10 mils (0.38 mm) may be used,although smaller values are already possible. However, it will beunderstood that as etching process capabilities improve, the width ofthe aperture body w may decrease. The aperture body width w may also beselected to other widths as known to those of skill in the art. In orderto minimize rearward facing radiation and maximize coupling to theresonant radiating cavity, the aperture may have a narrow aperture bodywidth, w, with wider openings at both aperture ends 868A and 868B. Inone embodiment, the aperture end width is more than five times widerthan the aperture body width. These aperture openings are describedfurther below.

In some embodiments, the shape of the aperture ends 868A and 868Bconsists of a combination of a rectangular end 872 and a semi-circularend 876. The rectangular end 872 has a width t and a length equal to theaperture end width W, and the semi-circular end 876 has a radius r. Insome embodiments, the radius r of the semi-circular end 876 is half ofthe aperture end width W. In some embodiments, the aperture end width Wis selected to be one third of the aperture length L, and the width t ofthe rectangular end 872 is one third of the radius r of thesemi-circular end 876 (or hence one sixth of the aperture end width W).Thus, the shape of the apertures is scalable according to a targetoperating frequency for the antenna assembly using the relationshipsbetween the aperture dimensions described in this paragraph (forexample, W=L/3, r=L/6, t=L/18), where a single variable, L, is scaledproportionally to the desired frequency of operation.

In one particular embodiment, the aperture length L is 10.42 mm for apatch array antenna assembly operating at 5300 MHz. In anotherparticular embodiment, the aperture length L is 9.98 mm for a patcharray antenna assembly operating at 5600 MHz. In yet another particularembodiment, the aperture length L is 9.7 mm for a patch array antennaassembly operating at 5788 MHz. It will be appreciated that the aperturemay have different aperture lengths L depending on, for example, theoperating frequency, the thickness of the spacer 508, and the size ofthe conducting patch element 604, as understood by those of skill in theart. It follows that ability to dynamically alter the electrical size ofthe aperture, whether by electrical or mechanical mechanism, allowsdynamic adjustment of the resonant frequency (or hence, the optimaloperating frequency) of the aperture feed structure and henceperformance of the antenna assembly.

In some embodiments, as shown in FIG. 8F, the projection of therespective pairs of apertures 820 and 824 within the first layer of thesecond substrate 512 located at the surface adjacent to spacer 508 uponthe outlines of the conductive patch elements 604 in the first substrate504. As can be seen from FIG. 8F, the apertures 820 and 824 do not needto be precisely centered relative to each conductive patch element 604.Each of the apertures 820 and 824 has a long axis parallel to the longdimension of the aperture and in the plane of the conductive groundplane, and a narrow axis parallel to the narrow dimension of theaperture and in the plane of the conductive ground plane. The magneticfields for the respective electromagnetic mode excited by an apertureare approximately constant inside the resonant radiating cavity in thedimension of the long axis, but vary significantly in the dimension ofthe narrow axis with maximum in the center of the conductive patchelement and diminishing to near zero at the edges of the conductivepatch element. The aperture electromagnetic coupling is primarilymagnetic coupling (inductive). Moving the apertures along the long axisdoes not significantly affect this magnetic coupling because themagnetic fields within the resonant radiating cavity are approximatelyconstant in this direction. Moving the apertures along the short axissignificantly affects the magnetic coupling mechanism because themagnetic fields vary in this direction. Hence, the arrangement ofapertures illustrated in FIG. 8F co-optimizes the respective magneticcoupling to the resonant radiating cavity in both apertures 820 and 824.Aperture 824 is shifted significantly along its long axis, and aperture820 is shifted slightly along its narrow axis.

FIG. 9 illustrates a diagram showing the surface current distribution ata pair of apertures 820, 824 showing additional advantages of theaperture design according to embodiments of the invention. In FIG. 9,feed lines 960 which terminate in a short circuit 828 drive the aperture820. As shown in FIG. 9, the current distortion owing to the secondorthogonal aperture 824 is minimized by rounded edges 928A and 928B,thereby making the input impedance of each aperture evenly balanced andimproving the radiation efficiency.

In an embodiment as described herein with four rows and two columns ofconductive patch elements operating at a target frequency of 5300 MHzand aperture dimensions as described above, a patch diameter is 28.5 mm,a conductive patch element thickness is 18 um, a spacer thickness is 2.4mm, a first substrate thickness is 0.508 mm, a second substratethickness is 0.762 mm, a center to center column spacing is 34 mm, acenter to center row spacing is 42.45 mm, outer dimensions for both afirst substrate and second substrate are 78 mm×185 mm, aperture 824offset is 5.5 mm from the center of the respective conductive patchelement and aperture 820 offset is 2.5 mm from the center of therespective conductive patch element. This embodiment achieves a patcharray antenna assembly with port to port isolation of >35 dB across theoperating band of 5250 MHz to 5350 MHz, a vertically polarized far-fieldradiation pattern with 16.3 dB gain, 16.1 degree vertical beamwidth, 42degree horizontal beamwidth, −0.8 dB radiation efficiency, and a secondhorizontally polarized far-field radiation pattern with 16.3 dB gain,16.9 degree vertical beamwidth, 39 degree horizontal beamwidth, and −0.8dB radiation efficiency.

FIG. 10A illustrates an exemplary unitary dipole antenna element 1000having a driven coplanar dipole 1032 with dipole branches 1032A and1032B and respective parasitic elements 1004A and 1004B. In oneembodiment, each unitary dipole antenna element 1000 will be arranged asan array of such elements to provide a horizontally polarized far-fielddirective gain antenna pattern. The structure of driven coplanar dipole1032 is referred to as coplanar because both dipole branches 1032A and1032B are located on the same surface of substrate 1012 as coplanar feedline pair 1008, which serve as the feed transmission line. Parasiticelements 1004A and 1004B broaden the radiation pattern in the plane ofthe surface of substrate 1012 having the driven coplanar dipole 1032 andthe coplanar feed line pair 1008, and may or may not be on the same sideof the substrate as driven coplanar dipole 1032 and coplanar feed linepair 1008.

Parasitic elements 1004A and 1004B are approximately half-wavelengthresonant dipole elements at the target operating frequency. In someembodiments, these parasitic elements 1004A and 1004B are asymmetricallyoffset towards the conductor connection end of substrate 1012 relativeto the axis of driven coplanar dipole 1032, as shown in FIG. 10A. Thisoffset enhances the mutual coupling between the driven coplanar dipolebranches 1032A and 1032B and respective parasitic elements 1004A and1004B. Driven coplanar dipole 1032 is approximately a one-halfwavelength resonant dipole and features strong electric fields at theends of dipole branches 1032A and 1032B where respective parasiticelements 1004A and 1004B are located. As a result, the couplingmechanism between driven coplanar dipole 1032 and parasitic elements1004 is primarily electric (capacitive), as opposed to magnetic(inductive). The electric fields created by driven coplanar dipole 1032are symmetric about the axis of dipole branches 1032A and 1032B. Thisrelative offset between driven coplanar dipole 1032 and parasiticelements 1004 causes the symmetric electrical fields of dipole branches1032A and 1032B to couple to a differential electromagnetic mode inrespective parasitic elements 1004A and 1004B. Half-wave dipolesresonate readily when excited via differential-mode electromagneticstimulus, but not to a common-mode electromagnetic stimulus, and so theoffset is necessary to achieve adequate mutual coupling. Adjusting thelength of the parasitic elements 1004A and 1004B controls the relativephase of the mutual coupling, and in turn the relative phase between theelectric current on driven coplanar dipole 1032 and the electric currenton parasitic elements 1004. Adjusting this relative phase between theseelectric currents achieves the desired far-field pattern. The inputimpedance of driven coplanar dipole 1032 can be tuned by adjusting thelength and shape of dipole branches 1032A and 1032B, whilst maintaininga fixed relative spacing to respective parasitic elements 1004A and1004B.

Coplanar feed line pair 1008 connects the driven unitary dipole antennaelement 1000. Dipoles are a balanced antenna, and as such are wellsuited to excitation by balanced transmission lines, such as coplanarstrips arranged as a coplanar feed line pair. Coplanar feed line pair1008 extends onto conductor connection substrate tab 1016. The conductorconnection substrate tab 1016 can be inserted into a slot (or conductorconnection cutout) on an orthogonal backplane substrate, where theincreased spacing 1020 between the branches of the coplanar feed linepair 1008 facilitates connection to another balanced transmission linestructure, such as coupled microstrip lines. In some embodiments, eachsubstrate 1012 having a unitary dipole antenna element 1000 is repeatedas individual elements in a dipole array antenna assembly, wherein eachindividual unitary dipole antenna element 1000 is coupled separately toa feed network on the orthogonal backplane substrate. Additionalfeatures for mechanical fastening to orthogonal substrates can beincluded, such as assembly slot 1024 and mechanical tabs 1028A and1028B. Each mechanical tab 1028A or 1028B can also have one or moremetalized pads 1034 as depicted in FIG. 10A that can be on either orboth surfaces of the substrate 1012 such that each mechanical tab alignswith an additional cutout in an orthogonal substrate and each metalizedpad 1034 adjoins another metalized pad on the orthogonal substrate forsoldering. These orthogonal substrates may include, but are not limitedto, an orthogonal backplane substrate as described herein. Metalized pad1022, typically located on the opposite surface of substrate 1012 fromthe surface comprising the coplanar feed line pair 1008, is an exemplaryfeature for providing additional mechanical fastening. The use of thesemechanical fastening features will be illustrated in further detailhereinafter.

In the embodiment depicted in FIG. 10A, a unitary dipole antenna elementfor a target operating frequency of 5.66 GHz comprises a driven coplanardipole 1032 with overall length of 18 mm and outer width of 1.5 mm, andparasitic elements 1004 of length of 16 mm and of outer width of 1.5 mm.The parasitic elements 1004 are located on the opposite side ofsubstrate 1012 from the driven coplanar dipole 1032 with rearward offsetof 1.5 mm relative to the centerline of the dipole branches 1032 andspacing of 11.05 mm from the center of coplanar strips to center ofparasitic elements. Coplanar feed line pair 1008 have an insideedge-edge spacing of 0.3 mm and each a width of 1 mm. The distance fromthe axial centerline of dipole branches 1032 to the bottom edge ofsubstrate 1012, the plane of the orthogonal backplane substrate, is 20mm.

FIG. 10B illustrates four coplanar dipole antenna elements 1060 each fedby a respective coplanar feed line pair 1056, and arranged as a verticalarray on common substrate 1036 to provide a vertically polarizedfar-field directive gain antenna pattern. Each coplanar dipole antennaelement 1060 comprises dipole branches 1060A and 1060B. For each ofcoplanar dipole antenna elements 1060, respective ones of coplanar feedline pair 1056 and dipole branches 1060A and 1060B are located on thesame surface of substrate 1036, forming a coplanar half-wavelengthresonant dipole. The coplanar feed line pair 1056 extends onto conductorconnection substrate tab 1040. The conductor connection substrate tab1040 can be inserted into a slot (or conductor connection cutout) on anorthogonal backplane substrate, where the increased spacing 1048 betweenthe branches of the coplanar feed line pair 1056 allows connection toanother balanced transmission line structure, such as coupled microstriplines. Additional features for mechanical fastening to orthogonalsubstrates can be included such as assembly slots 1052A to 1052E andmechanical tabs 1044A and 1044B. Each mechanical tab 1044A or 1044B canalso have one or more metalized pads 1046A and 1046B as depicted in FIG.10B that can be on either or both surfaces of the substrate 1036 suchthat each mechanical tab aligns with an additional cutout in anorthogonal substrate and each metalized pad 1034 adjoins anothermetalized pad on the orthogonal substrate for soldering. Theseorthogonal substrates may include, but are not limited to, an orthogonalbackplane substrate as described herein. Respective metalized pads 1058,typically located on the opposite surface of substrate 1036 fromcoplanar feed line pair 1056, are an exemplary feature for providingadditional mechanical fastening. The use of these mechanical fasteningfeatures will be illustrated in further detail hereinafter.

Coplanar dipoles antenna elements 1060 of FIG. 10B are all located onthe same surface of substrate 1036; however, in some embodiments, one ormore of coplanar dipole antenna elements 1060 may be located on one sideof the substrate 1036 and the remainder of the coplanar dipole antennaelements 1060 on the other side of substrate 1036.

In the embodiment depicted in FIG. 10B, coplanar dipole antenna elements1060 for a target operating frequency of 5.66 GHz have an overall lengthof 19.75 mm and outer width of 1.6 mm. Coplanar feed line pair 1056 hasan inside edge-to-edge spacing of 0.2 mm and each a width of 1 mm. Thecenter-to-center spacing between adjacent instances of coplanar dipoleantenna elements 1060 is 32.46 mm. The distance from the axialcenterline of dipole branches 1060 to the left side of substrate 1036,the plane of the orthogonal backplane substrate, is 21 mm.

FIG. 11A illustrates two instances of exemplary microstrip feedstructure portion 1100 of a transmission feed line network thatfacilitates electrical interconnect between a balanced element, such asone of unitary dipole antenna elements 1000 or coplanar dipole antennaelements 1060, to a feed network on orthogonal backplane substrate 1112.Two alternate orientations of the exemplary microstrip feed structureportion, 1100A and 1100B, are illustrated in FIG. 11A. In someembodiments, the backplane substrate 1112 is a printed circuit board.Orthogonal backplane substrate 1112 features a bottom outer layer (orfirst layer) that is a conductive plane 1144, which fulfills severalfunctions including providing a reflective plane for unitary dipoleantenna elements 1000 and coplanar dipole antenna elements 1060, andalso providing a ground plane for the microstrip feed structure portion1100.

Orthogonal backplane substrate 1112 contains a plurality of conductorconnection cutouts 1120 that are sized to accommodate conductorconnection substrate tabs 1040 and 1016. When conductor connectionsubstrate tab 1016, for example, is inserted into conductor connectioncutout 1120A, substrates 1012 and 1112 are oriented orthogonally to eachother. Three distinct connections can be made between the substrates1012 and 1112 using, for example, solder fillet. These connectionsinclude (1) one of the conductors that forms a branch of the coplanarfeed line pair 1008 is connected to microstrip line 1132A, (2) the otherconductor from the other branch of coplanar feed line pair 1008 isconnected to microstrip line 1108A, and (3) metalized pad 1022 isconnected to metalized pad 1136A for mechanical fastening (see also FIG.11B). Likewise, when conductor connection substrate tab 1040 is insertedinto conductor connection cutout 1120B that is rotated 90 degreesrelative to a conductor connection cutout 1120A, substrates 1036 and1112 are oriented orthogonally to each other. Again, three distinctconnections between substrates 1036 and 1112 can be made using, forexample, solder fillet. These three connections include: (1) one of theconductors that forms a branch of the coplanar feed line pair 1056 isconnected to microstrip line 1132B, (2) the other conductor from theother branch of coplanar feed line pair 1056 is connected to microstripline 1108B, and (3) metalized pad 1058 is connected to metalized pad1136B for mechanical fastening.

Conductive plane 1144 also has a respective conductor connectionclearance 1116A and 1116B to reduce parasitic capacitance betweencoplanar feed line pair 1008 or 1056 and the conductive plane 1144 inthe vicinity of the conductive junction for each conductor connectioncutout 1120. In one embodiment, conductor connection clearance 1116 isasymmetrically offset from conductor connection cutout 1120 as shown inFIG. 11A so as to be centered, both horizontally and vertically ineither orientation of conductor connection cutout 1120, about aprojected intersection of the coplanar feed line pair 1008 or 1056 withthe orthogonal substrate instead of being centered about the conductorconnection substrate tabs 1016 or 1040 (or centered about the conductorconnection cutouts 1120). This centering further minimizes parasiticcapacitance between the coplanar feed line pair 1008 or 1056 and theconductive plane 1144. For example, at a target operating frequency of5.66 GHz, the size of the conductor clearance is equivalent to thedistance between conductive plane 1144 and the plane that comprises thetransmission line feed structures such as microstrip feed structureportions 1100.

In some embodiments, the microstrip feed structure portion 1100 includesbalun elements to connect the balanced coplanar feed line pairs 1008 and1056 to the unbalanced microstrip lines 1104 and 1148 respectively.Microstrip lines 1104 and 1148 are both located within respectivetransmission feed line networks. Unbalanced microstrip lines are bettersuited for large parts of the transmission feed line networks as theyhave fewer conductors (and avoid cross-overs) compared to a balancedstructure such as coupled microstrip lines. However, at the actualelectrical connection point between the balanced coplanar feed line pair1008 or 1056 and its respective transmission feed line network, theconductive junction is preferably formed by a connection to a separatebalanced microstrip line from each branch of a coplanar feed line pairas described above. Thus, microstrip feed structure portion 1100 needsto at least include balanced microstrip lines at the conductive junctionand a balun structure that includes impedance matching between suchbalanced microstrip lines and unbalanced microstrip line 1104 within thetransmission feed line network. In some embodiments, the balun structureincludes the microstrip lines 1132 and 1108 and the T-junction 1124 asshown in FIG. 11A. In other embodiments (not shown), discrete componentsmay be used instead to transform balanced microstrip lines at theconductive junction to an unbalanced microstrip line 1104 within thetransmission feed line network as is known in the conventional art.

In the embodiment of FIG. 11A, the electrical length of microstrip line1132 is 90 degrees (or ¼ wavelength) at the desired operating frequency,and the electrical length of microstrip line 1108 is 270 degrees (or ¾wavelength) at the desired operating frequency. The bends in 1108provide for the additional electrical length required (180 degrees or ½wavelength) in a compact arrangement, and also provides a structure thatminimizes any distance where microstrip line 1108 runs parallel to otherparts of the same trace. Parallel lengths of microstrip line increaseundesired electrical coupling effects, which reduces the effectivenessof the balun structure. The T-junction 1124 provides a common connectionto both the 90 degree microstrip line 1132 and 270 degree microstripline 1108.

The feed structure 1100 could alternatively be implemented in astripline structure as opposed to a microstrip structure shown here iforthogonal backplane substrate 1112 is implemented as a multi-layer PCB.

FIG. 11B illustrates a cross-sectional view of an exemplary conductivejunction between orthogonal substrates 1012 and 1112. In someembodiments, as illustrated in FIG. 11B, the coplanar feed line pair1008, when assembled, can be easily soldered to the microstrip feedstructure portion 1100 to minimize overall feed losses and the metalizedpad 1022 can be easily soldered to the metalized pad 1136 to provideadditional mechanical ruggedness to the overall assembly.

FIG. 12 illustrates an equivalent electrical circuit representation ofmicrostrip feed structure portion 1100 for one embodiment. The loadimpedance, Z_(D) is the differential impedance of the coplanar feed linepair 1056 or 1008 at the plane where the connection is made tomicrostrip lines 1132 and 1108. In some embodiments, the driven unitarydipole antenna elements 1000 and driven coplanar dipole elements 1060are tuned for an input impedance of 100 ohms with the previouslydescribed spacing to reflective plane 1144, and the dimensions ofcoplanar feed line pair 1008 and 1056 are similarly chosen to achieve acharacteristic impedance of approximately 100 ohms. This tuning setsload impedance Z_(D) to approximately 100 ohms. The differentialimpedance is split evenly between the microstrip lines 1132 and 1108 atthe target operating frequency, such that each microstrip line has aneffective single-ended load impedance of about 50 ohms. In thisexemplary embodiment, each of the microstrip lines 1132 and 1108 has acharacteristic impedance of approximately 65 ohms. Microstrip lines 1132and 1108, both provide for quarter-wavelength impedance transformationand hence transform the 50 ohm load impedance to approximately 65̂2/50=85ohms, shown as Z_(A) in FIG. 12. The T-junction 1124 divides theimpedance Z_(A) by a factor of 2, owing to parallel impedances, makingZ_(B) equal to approximately 42 ohms. In some embodiments, an additionalmicrostrip line 1128 is included as a quarter-wavelength impedancetransformer, with characteristic impedance of approximately 65 ohms andtransforms the load impedance Z_(B) back to Z=100 ohms, which isconvenient for realizing transmission feed line networks for phasedarray applications. This circuit analysis is reciprocal and applieswhether the antenna is receiving or transmitting a signal.

Constraining the microstrip lines 1132 and 1108 to integer lengths ofquarter-wavelengths as described above integrates an impedance matchingfunction within the balun. The characteristic impedance of microstriplines 1132 and 1108 can be set to obtain the desired impedancetransformation ratio. In some embodiments, cascading a secondquarter-wavelength microstrip line 1128 allows the impedancetransformation to be spread over multiple elements, resulting in a morebroadband structure than choosing to do the required transformation inonly a single element, and provides flexibility to accommodate forvariations in element impedance without changing the other elements ofthe microstrip feed portion 1100.

When conductor connection substrate tabs 1040 and 1016 are inserted intorespective conductor connection cutouts 1120, coplanar feed line pair1008 or 1056 extends a short distance, such as 1 mm in some embodiments,beyond microstrip lines 1108 and 1132. This additional length isnecessary to provide adequately large surfaces for reliable solderjoints on each branch of coplanar feed line pair 1008 and 1056. Thisshort length of open circuit transmission line creates parasiticcapacitance at the conductive junction. The increased spacing 1048 and1029 results in parasitic inductance near the conductive junction. Inone embodiment, the length of the increased spacing 1048 and 1020 isoptimized such that the resulting parasitic inductance resonates withthe open stub parasitic capacitance, thereby adding no net additionalreactance at the conductive junction for the target operating frequency.For example, at a target operating frequency of 5.66 GHz, the length ofincreased spacing may be 1.25 mm.

FIGS. 13 and 14 illustrate an exemplary dipole array antenna assembly1304 based on an orthogonal assembly of five first substrates 1012A-E, asecond substrate 1036, and a third substrate 1300. Second substrate 1036contains four instances of coplanar dipole antenna elements 1060A-1060D.Here, the second substrate is used for vertical polarization if theentire antenna assembly 1304 is oriented in a backhaul radio, such asthe IBR. The longer dimension of assembly 1304 represents up/down andthe smaller dimension represents left/right. Each of the five firstsubstrates 1012A-E contains a unitary dipole antenna element 1000, usedhere for horizontal polarization given the antenna assembly orientationdescribed above. The third substrate 1300 is the orthogonal backplanesubstrate 1112 described above with a plurality of conductor connectioncutouts 1120, as well as various mechanical connection cutouts. In someembodiments, the third substrate 1300 is a multi-layer substrate havingat least two layers. In an exemplary embodiment, the first layer ofthird substrate 1300 comprises at least the conductive plane 1144 andthe respective conductor connection clearances 1116, as well as otherslots/clearances associated with mechanical tabs used for mechanicalassembly purposes. Also in an exemplary embodiment, the second layer ofthird substrate 1300 comprises at least the transmission line feednetworks including respective microstrip feed structure portions 1100for each element in the overall array. Thus, dipole array antennaassembly 1304 effectively interleaves vertically and horizontallypolarized antenna elements to create a two-port, orthogonally polarizeddipole array antenna assembly.

Although two-port, orthogonally polarized dipole array antennaassemblies with crossed dipole elements have been disclosed previously,these conventional antenna assemblies result in a crossed dipoleassembly that is more complex, costly, and prone to failure than thenovel interleaved array structure described as well as lower performingin terms of antenna efficiency and isolation. FIGS. 13A and 13B furtherillustrate the functionality of assembly slots 1024 and 1052, whichpermit the orthogonal, interleaved assembly of substrates 1012 and 1036.Upon assembly, second substrate 1036 effectively captures, retains, andprovides additional lateral support for the multiple instances of firstsubstrate 1012. Likewise, the multiple instances of first substrate 1012provide additional lateral support for substrate 1036. As shown in FIG.13A, each assembly slot 1052 aligns with a corresponding assembly slot1024 to set the spacing between successive first substrates in the arrayand to set the orientation of each first substrate 1012 as orthogonal tothe second substrate 1036. In some embodiments, the tabs can besoldered, both for mechanical retention and electrical connectivity, ina single soldering process, to corresponding pads or conductive feedlines on the second layer of the third substrate 1300.

The interleaved arrangement of the dipoles of opposite polarity alsoachieves very low mutual coupling between the elements of oppositepolarity. This is because the symmetric electric field of each element,couples in a common-mode fashion to the dipole elements that areorthogonally polarized. As previously discussed, half-wave dipoles tonot resonate in response to a common-mode excitation. There issignificant mutual coupling between the elements of a similarpolarization, but this coupling, which is deterministically known, canbe minimized through proper design of a feed network. It is important tominimize mutual coupling between orthogonally polarized elements, sothat the resulting orthogonally polarized antenna arrays do not couplesignificantly to each other. Mutual coupling between two arrays willreduce the efficiency of each antenna, and also has been shown toincrease correlation between the two antennas, resulting in degradedMIMO performance for a backhaul radio that uses such antenna assemblies.

FIG. 14 shows exemplary details of the third substrate 1300 (ororthogonal backplane substrate), which can be realized as a printedcircuit board. Third substrate 1300 features a first layer that is aconductive plane 1144, which fulfills several functions includingproviding a reflective plane for unitary dipole antenna elements 1000and coplanar dipole antenna elements 1060, and also providing a groundplane for the plurality of microstrip feed structure portions 1100 andthe microstrip distribution portions 1428A and 1428B of the firsttransmission line feed network 1408A and the microstrip distributionportions 1428C of the second transmission line feed network 1408B.Microstrip-based transmission line feed networks 1408A and 1408B connectto unitary dipole antenna elements 1000 and coplanar dipole antennaelements 1060A-1060D, respectively through a plurality of microstripfeed structure portions 1100 as detailed previously in FIGS. 11A and11B. Each microstrip-based transmission line feed network 1408A and1408B also comprises a respective feed point 1404A and 1404B andrespective microstrip distribution portions 1428A through 1428C as shownin FIG. 14. In some embodiments, additional components such as filtersand transmit power amplifiers or receive low noise amplifiers may belocated near the feed points 1404A and 1404B to minimize losses andimprove isolation performance in view of interconnects such as cablesfrom these dipole array antenna assemblies to the rest of the radio.

FIG. 14 also shows additional cutouts 1416 and metalized pads 1424 wherethe mechanical tabs 1028 and 1044 are fastened to the third substratepreferably in a single soldering step with the conductor connections tofurther increase mechanical rigidity and reliability of the overallantenna assembly. FIG. 14 further depicts that for one subset of five ofthe conductor connection cutouts, each conductor connection cutout 1120is arranged to align with the conductor connection substrate tabs 1016of the unitary dipole antenna elements, and that for the other subset offour of the conductor connection cutouts, each conductor connectioncutout 1120 is arranged to align with the conductor connection substratetabs 1040 of the coplanar dipole antenna elements. It will beappreciated that third substrate may have fewer than or more than fiveconductor connection cutouts in the vertical array subset and fewer thanor more than four conductor connection elements in the horizontal arraysubset.

Horizontally oriented dipoles tend to have a broader pattern beamwidthin elevation than the vertically oriented dipoles. The additional numberof elements in the horizontal array compared to the vertical array addsadditional array factor gain, such that the resulting elevationbeamwidth of the two polarizations is similar. FIG. 14 furtherillustrates the advantage of the compact size of microstrip feedstructure portion 1100, which permits both horizontal and verticaldipole array element interleaving at desirable element spacing, such as,for example 0.65 times the free-space wavelength at the target operatingfrequency, while leaving adequate room for the remaining transmissionfeed line network interconnects.

The transmission line feed network 1408B is a corporate feed networkproviding for uniform and matched group delay excitation of the fourvertically oriented coplanar dipole antenna elements 1060. Those skilledin the art will recognize that uniform excitation is commonly used todenote equal amplitude excitation amongst antenna elements, and that thematched group delay excites the elements in the same relative phase.Owing to the 2̂N number of elements (here N=2, or hence 4 elements),matched group delay from common feed point 1404B to each coplanar dipoleantenna element 1060, and uniform excitation of each coplanar dipoleantenna element 1060 is achieved via the symmetry of microstripdistribution portions 1428C. In this exemplary embodiment, thequarter-wavelength microstrip lines 1128 are adjusted differentlydepending on whether the associated coplanar dipole element 1060 is anouter antenna element in the array or an inner antenna element in thearray. This custom tuning is necessary to ensure the desired uniform,phase-aligned excitation of the plurality of coplanar dipole antennaelements 1060.

The transmission line feed network 1408A is also a uniformly excited,matched group delay corporate feed network, but the uneven number ofunitary dipole elements 1000 necessitates the inclusion of microstripdistribution portion 1428B which includes additional bends and lengthnecessary to achieve the desired matched group delay from common feedpoint 1404A to each unitary dipole element 1000. The quarter-wavelengthmicrostrip line 1128 that couples to microstrip distribution portion1428B is also uniquely shaped to accommodate the bends of microstripportion 1428B whilst also providing tuning for uniform excitation. Openstub tuning feature 1432 corrects for parasitic effects introduced inthe feed network owing to the undesired coupling between parallellengths of line and undesired parasitic capacitance at bends in themicrostrip distribution portions 1428A and 1428B. This open stub tuningfeature 1432 further ensures the desired uniform, phase-alignedexcitation of the plurality of unitary dipole antenna elements 1000.

Uniform and phase-aligned excitation of antenna elements in an arrayassembly is known to achieve the maximum realizable far-field directiveantenna gain pattern in the broadside direction; however, in someembodiments, non-uniform, and/or non-phase-aligned excitation may bepreferable. Both the amplitude and relative phase of each antennaelement can be adjusted to optimize some desired characteristic of theantenna array assembly, such as side lobe levels, peak gain orientation,and isolation to nearby antenna assemblies. These parameters may also bedynamically adjustable by including tuning elements within microstripdistribution portions 1428A, 1428B, and 1428C. In particular, adaptivecontrol of the phase, and or amplitude, of either each individualantenna element, or a subset of antenna elements, can be used todynamically tune both near-field and far-field coupling to an adjacentlylocated antenna assembly to achieve maximum port-to-port isolationbetween the two antenna assemblies.

In some applications it may be desirable to conform to an EIRP(effective isotropic radiated power) elevation mask, and in this case itis beneficial to shape the transmit antenna far-field radiation pattern,either statically or dynamically, to minimize the transmitted EIRP inthe vertical direction. This shaping can be achieved by altering therelative phase and/or the amplitude excitation of each antenna elementin the exemplary antenna array assemblies.

One exemplary embodiment for achieving far-field radiation patternshaping is by tapering the amplitude excitement of the individualantenna elements within the array assembly by some pattern as a functionof the location of the antenna element in the array assembly. A taperedamplitude excitation, wherein the inner antenna elements in an antennaarray assembly are driven with higher relative amplitude than the outerantenna elements, is known to achieve lower far-field antenna patternside lobe levels than the equivalent array assembly with uniformexcitation. Microstrip lines 1140A to 1140B and 1128A to 1128B can beused to control the input impedances of each element in the dipole arrayantenna assembly as seen by transmission line feed networks 1408A and1408B. This impedance can be adjusted to control the relative amplitudeexcitation of the elements, providing the desired tapered amplitudeexcitation, and in turn, the desired far-field antenna radiation patternside lobe suppression. Similarly, these amplitude tapers can be appliedto the aperture-fed patch element array described herein.

Another exemplary embodiment for far-field antenna radiation patternshaping is via a progressive relative phase shift in the relativeexcitation of each antenna element in the array antenna assembly. Aprogressive relative phase shift between the antenna elements of anantenna array assembly is known to scan the main beam of the far-fieldantenna radiation pattern in an angular sense. The lengths of microstripline distribution portions 1428 can be adjusted to vary the relativephase excitation of each element. Alternatively, electronic phaseshifters can be inserted into microstrip line distribution portions 1428to dynamically vary the individual antenna element relative phaseexcitation. The ability to dynamically scan the main beam in a downwarddirection can help conform to the EIRP elevation mask in response tochanges in elevation alignment of the IBR. Similarly, these progressiverelative phase shifts can be applied to the aperture-fed patch elementarray described herein.

For example, if the IBR is installed with a tilt angle upwards towardsthe sky then a sensor such as based on a multi-axis accelerometer candetermine the amount of upward tilt and then a controller, such as theRRC, can provide or cause to be provided certain control signals to theantenna array assembly, whether based on aperture-fed patch elements orsubstrate tab connected dipole elements or otherwise, so that the mainbeam is either adjusted in a downward direction or has additionalsidelobe suppression applied, thereby either optimizing link performanceand/or conforming with a regulatory domain elevation mask EIRP limit ata particular elevation angle.

In one embodiment, the use of progressive relative phase shifts and/oramplitude tapers between either aperture-fed patch elements or substratetab connected dipole elements that can be dynamically altered is appliedbased on a tilt sensor input to ensure that the maximum EIRP above anupward elevation such as 30 degrees or higher is at least 13 dB lowerthan the maximum EIRP at zero degrees elevation angle, or alternativelyto ensure that the maximum EIRP above an upward elevation such as 30degrees or higher is not greater than a prescribed limit such as +23dBm. In other embodiments, the IBR uses the tilt sensor input and theknown characteristics of a particular antenna assembly far-fieldradiation pattern to limit the maximum conducted power into the antennaassembly to ensure that the maximum EIRP above an upward elevation suchas 30 degrees or higher is not greater than a prescribed limit such as+23 dBm.

In the example described herein with five elements in the horizontalarray and four elements in the vertical array operating at a targetfrequency of 5660 MHz and with the dipole element dimensions describedabove, the dipole array antenna assembly achieves port-to-port isolationof more than 35 dB across the operating band of 5470 MHz to 5850 MHz, avertically polarized far-field radiation pattern with 11.1 dB gain, 19.5degree vertical beamwidth, 120.1 degree horizontal beamwidth, −0.75 dBradiation efficiency, and a second horizontally polarized far-fieldradiation pattern with 11.5 dB gain, 16.7 degree vertical beamwidth,122.7 degree horizontal beamwidth, and −0.75 dB radiation efficiency.

Numerous additional variations of the above-described elements of theIBR and antennas can also be advantageously utilized in substitution foror in combination with the exemplary embodiments described above. Forexample, in certain embodiments the aperture-fed patch array antennaassemblies are used as directive gain antenna elements that can becoupled to receive RF chains and the dipole array antenna assemblies areused as directive gain antenna elements that can be coupled to transmitRF chains. In other exemplary embodiments, the aperture-fed patch arrayantenna assemblies are used as directive gain antenna elements that canbe coupled to both receive RF chains and transmit RF chains, typicallywherein a first subset of such antenna assemblies is configured forreceive usage and a second subset is configured for transmit usage. Whenan aperture-fed patch array antenna assembly is configured for transmitusage, active components such as power amplifiers and filters may alsobe integrated into such antenna assemblies preferably with minimal lossbetween the feed points and the power amplifiers.

One or more of the methodologies or functions described herein may beembodied in a computer-readable medium on which is stored one or moresets of instructions (e.g., software). The software may reside,completely or at least partially, within memory and/or within aprocessor during execution thereof. The software may further betransmitted or received over a network.

The term “computer-readable medium” should be taken to include a singlemedium or multiple media that store the one or more sets ofinstructions. The term “computer-readable medium” shall also be taken toinclude any medium that is capable of storing, encoding or carrying aset of instructions for execution by a machine and that cause a machineto perform any one or more of the methodologies of the presentinvention. The term “computer-readable medium” shall accordingly betaken to include, but not be limited to, solid-state memories, andoptical and magnetic media.

Embodiments of the invention have been described through functionalmodules at times, which are defined by executable instructions recordedon computer readable media which cause a computer, microprocessors orchipsets to perform method steps when executed. The modules have beensegregated by function for the sake of clarity. However, it should beunderstood that the modules need not correspond to discrete blocks ofcode and the described functions can be carried out by the execution ofvarious code portions stored on various media and executed at varioustimes.

It should be understood that processes and techniques described hereinare not inherently related to any particular apparatus and may beimplemented by any suitable combination of components. Further, varioustypes of general purpose devices may be used in accordance with theteachings described herein. It may also prove advantageous to constructspecialized apparatus to perform the method steps described herein. Theinvention has been described in relation to particular examples, whichare intended in all respects to be illustrative rather than restrictive.Those skilled in the art will appreciate that many differentcombinations of hardware, software, and firmware will be suitable forpracticing the present invention. Various aspects and/or components ofthe described embodiments may be used singly or in any combination. Itis intended that the specification and examples be considered asexemplary only, with a true scope and spirit of the invention beingindicated by the claims.

What is claimed is:
 1. An antenna assembly comprising: a plurality offirst substrates each comprising a unitary dipole antenna element,wherein each unitary dipole antenna element comprises a first pair ofdipole branches, a first coplanar feed line pair and a first conductorconnection substrate tab; a second substrate comprising a plurality ofcoplanar dipole antenna elements, wherein each coplanar dipole antennaelement comprises a second pair of dipole branches, a second coplanarfeed line pair and a second conductor connection substrate tab; and athird substrate comprising a plurality of conductor connection cutouts,a first layer and a second layer, wherein the first layer comprises aconductive plane with a plurality of conductor connection clearances andwherein the second layer comprises a first transmission line feednetwork and a second transmission line feed network; wherein the secondsubstrate is orthogonal to each of the plurality of first substrates andwherein the third substrate is orthogonal to the second substrate andeach of the plurality of first substrates; wherein the firsttransmission line feed network conductively connects to each respectiveunitary dipole antenna element via its respective first coplanar feedline pair at a respective one of a plurality of first conductivejunctions, each said first conductive junction comprising the respectivefirst conductor connection substrate tab, a first corresponding one ofthe plurality of conductor connection cutouts, and a first correspondingone of the plurality of conductor connection clearances; and wherein thesecond transmission line feed network conductively connects to eachrespective coplanar dipole antenna element via its respective secondcoplanar feed line pair at a respective one of a plurality of secondconductive junctions, each said second conductive junction comprisingthe respective second conductor connection substrate tab, a secondcorresponding one of the plurality of conductor connection cutouts, anda second corresponding one of the plurality of conductor connectionclearances.
 2. The antenna assembly of claim 1, wherein the first pairof dipole branches of each unitary dipole antenna element is located ona same surface as the first coplanar feed line pair.
 3. The antennaassembly of claim 1, wherein each unitary dipole antenna element furthercomprises a first pair of parasitic elements.
 4. The antenna assembly ofclaim 3, wherein the first pair of parasitic elements of each unitarydipole antenna element is located on the same surface as the first pairof dipole branches.
 5. The antenna assembly of claim 3, wherein thefirst pair of parasitic elements broadens a radiation pattern of eachunitary dipole antenna element in a plane of the same surface as thefirst pair of dipole branches.
 6. The antenna assembly of claim 3,wherein the first pair of parasitic elements comprise half-wavelengthresonant dipole elements at a target operating frequency of the antennaassembly.
 7. The antenna assembly of claim 3, wherein the first pair ofparasitic elements are asymmetrically offset relative to an axis of therespective first pair of dipole branches towards an end of therespective first substrate comprising the respective first conductorconnection substrate tab.
 8. The antenna assembly of claim 1, whereineach of the plurality of first substrates further comprises a firstassembly slot and wherein the second substrate further comprises aplurality of second assembly slots.
 9. The antenna assembly of claim 8,wherein a respective one of the plurality of second assembly slotsaligns with a respective first assembly slot within each respectivefirst substrate.
 10. The antenna assembly of claim 1, wherein each ofthe plurality of first substrates further comprises one or more firstmechanical tabs.
 11. The antenna assembly of claim 10, wherein the thirdsubstrate further comprises additional cutouts, each additional cutoutcorresponding to a respective first mechanical tab amongst the pluralityof first substrates.
 12. The antenna assembly of claim 11, wherein eachof the plurality of first substrates further comprises one or more firstmetalized pads corresponding to respective ones of each first mechanicaltab, wherein the second layer of the third substrate further comprises aplurality of third metalized pads corresponding to respective ones ofeach first mechanical tab, and wherein each first metalized pad adjoinsa respective third metalized pad.
 13. The antenna assembly of claim 1,wherein the second substrate further comprises one or more secondmechanical tabs.
 14. The antenna assembly of claim 13, wherein the thirdsubstrate further comprises additional cutouts, each additional cutoutcorresponding to a respective second mechanical tab.
 15. The antennaassembly of claim 14, wherein the second substrate further comprises oneor more second metalized pads corresponding to respective ones of eachsecond mechanical tab, wherein the second layer of the third substratefurther comprises a plurality of third metalized pads corresponding torespective ones of each second mechanical tab, and wherein each secondmetalized pad adjoins a respective third metalized pad.
 16. The antennaassembly of claim 1, wherein each of the plurality of conductorconnection clearances is asymmetrically offset relative to a respectiveone of the plurality of conductor connection cutouts.
 17. The antennaassembly of claim 6, wherein said asymmetric offset centers each of theplurality of conductor connection clearances relative to a projectedintersection with the third substrate for a respective one of firstcoplanar feed line pairs or second coplanar feed line pairs.
 18. Theantenna assembly of claim 11, wherein the second substrate is orientedsuch that each of the plurality of coplanar dipole antenna elementsradiates in a vertical polarization far-field pattern and wherein theplurality of first substrates are oriented such that each unitary dipoleantenna element radiates in a horizontal polarization far-field pattern.19. The antenna assembly of claim 1, wherein the first transmission linefeed network comprises a first feed point, a first microstripdistribution portion, and a plurality of first microstrip feed structureportions and the second transmission line feed network comprises asecond feed point, a second microstrip distribution portion, and aplurality of second microstrip feed structure portions.
 20. The antennaassembly of claim 19, wherein each first microstrip feed structureportion comprises a first balun structure that couples a first pair ofbalanced microstrip lines at a respective one of the plurality of firstconductive junctions to a first unbalanced microstrip line within thefirst microstrip distribution portion and each second microstrip feedstructure portion comprises a second balun structure that couples asecond pair of balanced microstrip lines at a respective one of theplurality of second conductive junctions to a second unbalancedmicrostrip line within the second microstrip distribution portion. 21.The antenna assembly of claim 20, wherein each of the first and secondbalun structures comprises a first microstrip line, a second microstripline, and a T-junction, and wherein the second microstrip line iselectrically longer than the first microstrip line by one halfwavelength at a target operating frequency of the antenna assembly andthe second microstrip line comprises at least one additional bend thanthe first microstrip line.
 22. The antenna assembly of claim 21, whereineach of the first and second microstrip lines functions as an impedancetransformer of an electrical length that is an integer multiple of onequarter wavelength at a target operating frequency of the antennaassembly.
 23. The antenna assembly of claim 21, wherein each of thefirst microstrip feed structure portion and the second microstrip feedstructure portion further comprises an impedance transformer from theT-junction within its respective first or second balun structure to itsrespective first or second unbalanced microstrip line within therespective first or second microstrip distribution portion.
 24. Theantenna assembly of claim 23, wherein the impedance transformercomprises an unbalanced microstrip line of an electrical length that isan integer multiple of one quarter wavelength at a target operatingfrequency of the antenna assembly.
 25. The antenna assembly of claim 19,wherein the first feed point and the second feed point are each coupledto respective components on the second layer of the third substrate. 26.The antenna assembly of claim 25, wherein the respective components areat least one of an RF filter or a power amplifier within a transmitter.27. The antenna assembly of claim 19, wherein the first microstripdistribution portion equally divides a first power and matches a firstgroup delay from the first feed point to each of the plurality of firstmicrostrip feed structure portions and the second microstripdistribution portion equally divides a second power and matches a secondgroup delay from the second feed point to each of the plurality ofsecond microstrip feed structure portions.
 28. The antenna assembly ofclaim 19, wherein each of the first microstrip distribution portion andthe second microstrip distribution portion comprises at least onetunable element.
 29. The antenna assembly of claim 28, wherein an inputsignal applied to at least one tunable element adjusts at least onecharacteristic of the antenna assembly, said characteristic being one ormore of a far-field radiation pattern, a coupling between the first feedpoint and the second feed point, or a coupling to one or more nearbyantennas.
 30. The antenna assembly of claim 1, wherein a numerical countof unitary dipole antenna elements exceeds that of a numerical count ofcoplanar dipole antenna elements.